Semiconductor integrated circuit device with a plurality of logic circuits having active pull-down functions

ABSTRACT

In accordance with one aspect of the invention, a semiconductor integrated circuit is provided wherein an input circuit is formed by a phase split circuit having a bipolar transistor which outputs an inverted output from the collector and a non-inverted output from the emitter. The emitter follower output circuit is driven by an inverted output of the phase split circuit. Meanwhile, an emitter load of the emitter follower output circuit is formed by a transistor, and the emitter load transistor is temporarily driven conductively by a charging current of the capacitance to be charged by the rising edge of the non-inverted output of the phase split circuit. As a second aspect of the invention, a logic circuit is formed of a logic portion and an output portion. The output portion includes an emitter follower output transistor receiving an output signal generated by the logic portion and an active pull-down transistor receiving at its base a signal supplied thereto through a capacitance element. The signal received by the active pull-down transistor has a phase reverse to that of the input signal supplied to the base of said output transistor. Between the base and emitter of the active pull-down transistor, there is disposed a bias circuit formed of a transistor receiving at its base a predetermined bias voltage and an emitter resistor. Further, between a junction point of the emitter follower output transistor and the active pull-down transistor and the emitter of the transistor as a constituent of the bias circuit, there is disposed a capacitance element for feeding back the output signal.

This is a continuation of application Ser. No. 557,109, filed Jul. 25,1990; which is a continuation-in-part application of Ser. No. 330,461filed on Mar. 30, 1989, now U.S. Pat. No. 4,999,520.

BACKGROUND OF THE INVENTION

The present invention relates to a logic circuit, particularly to theart effectively applied to a high speed logic circuit to be formedwithin a semiconductor integrated circuit, namely to the art effectivelyapplied, for example, to NTL (Non-Threshold Logic), or ECL (EmitterCoupled Logic) and, moreover, to the art effectively applied to thebasic logic circuit within a bipolar gate array integrated circuit (forexample, refer to Nikkei Electronics, No. 420, pp. 117-120, May 4, 1987,Nikkei McGraw-Hill, Inc.).

The present invention also relates to a technique effective for use inan NTL or ECL circuit provided with an active pull-down circuit.

Some NTL and ECL circuits proposed receive low amplitude digital inputsignals and conduct high speed logic operations. Moreover, an NTLcircuit has been proposed with an output emitter follower and an ECLcircuit with an emitter follower formed by adding the output emitterfollower circuit to an NTL circuit and an ECL circuit (hereinafter, theNTL circuit with an output emitter follower is called an NTL circuit andthe ECL circuit with an output emitter follower is called an ECLcircuit, respectively), a high speed logic integrated circuit comprisingthe basic configuration of the NTL circuit and the high speed logiccircuit comprising the basic configuration of an ECL circuit and an NTLcircuit.

FIG. 32 is a configuration example of the logic circuit of prior art.

The logic circuit LOG₇ shown in the same figure is formed as an NTLcomprising a first transistor Q₃₆ forming a grounded emitter type phaseinversion circuit and a second transistor Q₃₇ forming an emitterfollower output circuit, and a negative logic output V₀₁₀ can beobtained for an input V_(i7) as shown in FIG. 33 by applying aninversion output extracted from the collector of transistor Q₃₆ to thebase of transistor Q₃₇.

In this case, this logic circuit LOG₇ is provided with a collector loadresistance R₃₄, an emitter bias resistance R₃₅, a speedup capacitanceC_(a8) for improving switching operation of transistor Q₃₆ and anemitter load resistance R₃₆ of transistor Q₃₇.

The NTL circuit is described, for example, in Japanese Patent Laid-OpenNo. 63-124615.

SUMMARY OF THE INVENTION

The inventors of the present invention, however, have found that the NTLcircuit technique described above is accompanied by the followingproblems.

Namely, as the operation waveforms of logic circuit LOG₇ described areshown in FIG. 33, when the input V_(i7) is changed to low level L fromhigh level H and thereby the output V₀₁₀ is changed to the high level Hfrom the low level L, a load capacitance CL₃ is actively charged by theemitter follower operation of transistor Q₃₇ and thereby the output V₀₁₀can immediately be boosted up to the high level H from the low level L.

However, when the input V_(i7) is changed to the high level H from thelow level L and thereby the output V₀₁₀ is changed to the low level Lfrom the high level H, it is required to wait for the charge of loadcapacitance CL₃ to be passively discharged by the emitter loadresistance R₃₆. Therefore, a comparatively longer delay time t_(pd2) isgenerated at the change of output V₀₁₀ to the low level L from the highlevel H.

In above logic circuit LOG₇, it is necessary for ensuring a shortfalling time of output V₀₁₀ to sufficiently lower the resistance valueof emitter load resistance R₃₆ of the transistor Q₃₇ to cause adischarge current I₉ from the load capacitance CL₃ to easily flow.However, when the value of emitter load resistance R₃₆ is lowered, acurrent steadily flowing into the emitter load resistance R₃₆ when theoutput V₀₁₀ is high level H increases, also resulting in increase ofcurrent dissipation of the circuit.

As described above, the logic circuit described has a problem that it isdifficult to simultaneously satisfy the low power consumption and highspeed operation.

Moreover, a problem also rises that if the circuit elements aresuperminiaturized by the monolithic semiconductor integrated circuittechnology since the logic circuit described, for example, is used asthe basic logic circuit of a large capacity gate array integratedcircuit, malfunction may easily be generated due to the influence ofα-rays.

For the logic circuit formed by ECL, the improved circuit is indicatedin the U.S. Pat. No. 4,539,493.

It is therefore an object of the present invention to provide asemiconductor integrated circuit which simultaneously satisfies highspeed characteristic and low power consumption Another object of thepresent invention is to provide a semiconductor integrated circuit whichis resistive to the influence of α-rays.

These and other objects and novel features of the present invention willbe become apparent from the following description of the specificationand the accompanying drawings of the present invention.

A typical disclosure of the present invention will be summarizedhereinafter.

Namely, in the logic circuit forming NTL, the input circuit is formed bya phase split circuit, the emitter follower output circuit is drivenwith an inverted output of this phase split circuit, on the other hand,an emitter load of emitter follower output circuit is formed by atransistor and the emitter load transistor is temporarily conductivelydriven with a charging current of the capacitance which is charged bythe rising edge of a non-inverted output of such phase split circuit.

According to the means described above, in the transition period whereinthe emitter follower output circuit transistor is changed to thenon-conductive from the conductive condition at the transition of inputlogic, the emitter load transistor of the emitter follower outputcircuit temporarily becomes conductive quickly discharging the loadcapacitance, and in the steady state other than the transition period,the emitter load transistor keeps an almost non-conductive state.

Thereby, the objects for simultaneously satisfying low power consumptionand high speed characteristics, enabling the wired logic and receivinglittle influence from α-rays can be attained.

Moreover, according to investigations by the inventors of the presentinvention, the current circuit speed of the NTL circuit is 70 ps/gate(under the loading condition) but it may be improved in the future up to30 ps/gate by applying the present invention to the logic circuit,namely the NTL circuit of FIG. 32. In addition, the power consumption ofthis NTL circuit may be lowered up to 1/2 to 1/5 in comparison with theECL circuit of the prior art.

Furthermore, according to investigations by the inventors of the presentinvention, a logic circuit to which the present invention is applied isa low amplitude circuit and therefore realizes high speed operation. Inaddition, the power supply voltage of the logic circuit (logic part) towhich the present invention is applied is a low voltage circuit (lowvoltage part) allowing a voltage as low as -2.0--1.2 V and thereforerealizes low power supply voltage.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 illustrates the first principle of a first logic circuit to whichthe present invention is applied;

FIG. 2 is an input/output characteristic diagram of the first logiccircuit of FIG. 1;

FIG. 3 illustrates waveforms operation examples of the first logiccircuit of FIG. 1;

FIG. 4 is a sectional view of bipolar transistor and collector loadresistance included in the first logic circuit shown in FIG. 1;

FIG. 5 is a first circuit diagram for realizing the first principlediagram of the first logic circuit shown in FIG. 1;

FIG. 6 is a characteristic graph indicating relationship between basevoltage and collector current of a pull-down transistor included in thefirst logic circuit shown in FIG. 5;

FIG. 7A illustrates waveforms indicating the simulation of the input andoutput signals of the first logic circuit shown in FIG. 5;

FIG. 7B illustrates waveforms indicating relationship between collectorcurrent of emitter follower transistor and collector current of pulldown transistor included in the first logic circuit shown in FIG. 5during the simulation operation;

FIG. 8 is a circuit diagram providing a plurality of differentialcapacitances in the first logic circuit shown in FIG. 5;

FIG. 9 is a layout diagram indicating a plurality of differentialcapacitances included in the first logic circuit shown in FIG. 8;

FIG. 10 is a circuit diagram indicating application example of the firstlogic circuit shown in FIG. 5;

FIG. 11 is a circuit diagram indicating the first application example ofthe first logic circuit shown in FIG. 10;

FIG. 12 is a circuit diagram indicating the second application exampleof the first logic circuit shown in FIG. 10;

FIG. 13 is a circuit diagram coupling the first logic circuit shown inFIG. 10 to another logic circuit;

FIG. 14 is a circuit diagram stabilizing an output of the first logiccircuit shown in FIG. 10;

FIG. 15 is a layout of the first logic circuit shown in FIG. 14;

FIG. 16 is a second circuit diagram for realizing the first principlediagram of the first logic circuit shown in FIG. 1;

FIG. 17 is a circuit diagram indicating an application example of thefirst logic circuit shown in FIG. 16;

FIG. 18 is a third circuit diagram for realizing the first principlediagram of the first logic circuit shown in FIG. 1;

FIG. 19 is a circuit diagram providing a constant voltage generatingcircuit in the first logic circuit shown in FIG. 5;

FIG. 20 is a circuit diagram indicating the second principle diagram ofthe first logic circuit to which the present invention is applied;

FIG. 21 illustrates waveforms indicating operation example of the firstlogic circuit shown in FIG. 20;

FIG. 22 is a circuit diagram for realizing the second principle diagramof the first logic circuit shown in FIG. 20;

FIG. 23 is a circuit diagram indicating the second logic circuit towhich the present invention is applied;

FIG. 24 is a circuit diagram indicating the third logic circuit to whichthe present invention is applied;

FIG. 25 is a circuit diagram indicating a fourth logic circuit to whichthe present invention is applied;

FIG. 26 is a circuit diagram indicating the fifth logic circuit to whichthe present invention is applied;

FIG. 27 is a schematic plan view indicating a large capacity gate arrayintegrated circuit to which the present invention is applied;

FIG. 28 is a circuit diagram forming a cell of a large capacity gatearray integrated circuit shown in FIG. 27 with the first logic circuit;

FIG. 29 is a sectional view of a large capacity gate array shown in FIG.28 sealed by microchip carrier;

FIG. 30 is a sectional view forming a capacitance chip on the carriershown in FIG. 29;

FIG. 31 is diagram indicating mounting of a microchip carrier shown inFIG. 29 on a mounting substrate;

FIG. 32 is a circuit diagram indicating an configuration example of thelogic circuit discussed prior to the present invention;

FIG. 33 illustrates waveforms indicating operation example of the logiccircuit shown in FIG. 32;

FIG. 34 is a circuit diagram showing an embodiment in which an activepull-down circuit according to the present invention is applied to anNTL circuit;

FIG. 35 is a circuit diagram showing another embodiment in which anactive pull-down circuit according to the present invention is appliedto an NTL circuit;

FIG. 36 is a circuit diagram showing a further embodiment in which anactive pull-down circuit according to the present invention is appliedto an NTL circuit;

FIG. 37 is a circuit diagram showing an embodiment in which an activepull-down circuit according to the present invention is applied to anECL circuit;

FIG. 38 is a circuit diagram showing another embodiment in which anactive pull-down circuit according to the present invention is appliedto an NTL circuit; and

FIG. 39 is a circuit diagram showing an example of an active pull-downcircuit investigated by the present inventor in the process ofaccomplishing the present invention.

DESCRIPTION OF THE PREFERRED EMBODIMENTS

FIG. 1 illustrates a first principle diagram of the first logic circuitto which the present invention is applied. In the following figures, thelike elements are given the like numerals, a bipolar transistor shown isan NPN transistor, and logic circuits shown in the figures form,although not limited particularly,.the one of cells as the basic logiccircuit of the large capacity gate array integrated circuit as shown inFIG. 27 described later, which is formed, together with the other cells,on a single P type semiconductor substrate such as a single crystalsilicon. In FIG. 1, the logic circuit LOG₁ is formed by a phase splitcircuit 1, an emitter follower output circuit 2 which is driven by aninverted output -V_(i) of this phase split circuit 1, an activepull-down circuit 3, and a differential capacitance C_(al) whichdifferentiates the rising edge of the non-inverted output +V_(i) of thephase split circuit 1.

In this case, the phase split circuit 1 described is formed by a bipolartransistor Q₁, a collector load resistance R₂ located in series betweenthe collector of such bipolar transistor Q₁ and a high level powersource voltage V_(cc) and an emitter load resistance R₃ located inseries between the emitter of bipolar transistor Q₁ and a low levelpower source voltage V_(ee).

The emitter follower output circuit 2 is formed by an emitter followertransistor Q₂. The emitter of emitter follower transistor Q₂ isconnected to the low level power source voltage V_(ee) through thepull-down transistor Q₃ described later and simultaneously with a loadcapacitance CL₁ generated equitably by distributed capacitance of anoutput lead wire and input capacitance of logic circuit in the nextstage.

The active pull-down circuit 3 is formed by the pull-down transistor Q₃,which is an emitter load of the emitter follower output circuit 2 andsuch differential capacitance C_(al) is conductively driven temporarilyby differentiating the rising edge of non-inverted output +V_(i).

The differential capacitance C_(al) is connected, at the one electrode,with the emitter of bipolar transistor Q₁ and, at the other electrode,with the base of pull-down transistor Q₃. The base of pull-downtransistor Q₃ is connected in parallel with a resistance R₁. Thisresistance R₁ forms, together with the differential capacitance C_(al),a differential time constant and simultaneously a discharge path for theremaining charge of the base of pull-down transistor Q₃.

To the base of bipolar transistor Q₁ forming the phase split circuit 1,the digital input signal V_(i1) is supplied from the other logiccircuits not illustrated in the large capacity gate array integratedcircuit. Here, the power source voltage V_(cc) of circuit is set to theground potential, while the power source Voltage V_(ee) of circuit isset to a predetermined negative power source voltage (for example, -2V). Moreover, the digital input signal V_(i1) is given comparativelysmall signal amplitude, for example, with the high level V_(H) thereofset to -0.8 V or with the low level V_(L) to -1.4 V.

FIG. 2 is an input/output characteristic graph the logic circuit LOG₁shown in FIG. 1. In FIG. 2, the level of digital output signal V_(o1) ofthe logic circuit LOG₁ shown in FIG. 1 will be explained.

When the digital input signal V_(i1) is set to the predetermined lowlevel V_(L), the collector current I_(cl) of bipolar transistor Q₁becomes a comparatively small value as indicated below when the baseemitter voltage of bipolar transistor Q₁ and current conduction rate arerespectively designated as V_(BE1) and α₁.

    I.sub.cl =(V.sub.L -V.sub.BE1 -V.sub.ee)×α.sub.1 /α.sub.3

In this case, a voltage of node n₁ is set to such a high level as

    V.sub.V.sub.HC =-R.sub.2 ×Ic.sub.1

The high level V_(HC) of node n₁ is further shifted as much as baseemitter voltage V_(BE2) of transistor Q₂ forming the output emitterfollower circuit. Thereafter, the digital output signal V_(o1) of thelogic circuit LOG₁ is set to the high level as indicated below.

    V.sub.H =V.sub.HC -V.sub.BE2

The high level V_(H) of digital output signal V_(o1) is set to the highlevel V_(H) of digital input signal by setting the collector loadresistance R₂ and emitter load resistance R₃.

Meanwhile, the digital input signal V_(o1) is set to the predeterminedhigh level V_(H), the collector current I_(Cl), of bipolar transistor Q₁becomes a comparatively large value as indicated below when the baseemitter voltage of bipolar transistor Q₁ and current conduction rate aredefined respectively as V_(BE1), and α_(1').

    I.sub.cl' =(V.sub.H -V.sub.BE1' -V.sub.ee)×α.sub.1' /R.sub.3

In this case, a voltage of node n₁ is set to a level as low as

    V.sub.LC =-R.sub.2 ×I.sub.cl'

The low level V_(LC) of node n₁ is set to the low level of digitaloutput signal V_(o1) of the logic circuit LOG₁ after it is shifted asmuch as the base emitter voltage V_(BE2), of the emitter followertransistor Q₂. In this case, the low level V_(L) of digital outputsignal V_(o1) is indicated as

    V.sub.L =V.sub.LC -V.sub.BE2'

The low level V_(L) of the digital output signal V_(o1) can be set tothe low level V_(L) of digital input signal V_(i1) by setting thecollector load resistance R₂ and emitter load resistance R₃.

FIG. 3 illustrates waveforms indicating an operation example of thelogic circuit LOG₁ shown in FIG. 1.

In FIG. 3, potentials of digital input signal V_(i1) and digital outputsignal V_(o1) are plotted on the vertical axis, while the time is shownon the horizontal axis.

As indicated in the same figure, when the digital input signal V_(i1)changes to the low level V_(L) from the high level V_(H), the loadcapacitance CL₁ is quickly charged by the emitter follower operation ofthe emitter follower transistor Q₂. Thereby, the digital output signalV_(o1) immediately rises up to the high level V_(H) from the low levelV_(L) as in the case of the ECL circuit of the prior art.

When the digital input signal V_(i1) changes to the high level V_(H)from the low level V_(L), a charging current I_(l) of the differentialcapacitance C_(al) flows toward the base of pull-down transistor Q₃ fromthe emitter of bipolar transistor Q₁. Thereby, the pull-down transistorQ₃ is temporarily conductively driven, quickly discharging the loadcapacitance CL₁. Namely, the differential capacitance C_(al)differentiates the rising edge of non-inverted output +V_(i) of thephase split circuit 1 and this differential output temporarilyconductively drives the pull-down transistor Q₃. As a result, a largedischarge current I₂ is extracted from the load capacitance CL₁ and thedigital output signal V_(o1) immediately drops to the low level V_(L)from the high level V_(H).

As described above, a delay time t_(pd1) when the digital output signalV_(o1) drops is sharply curtailed in the logic circuit LOG₁ describedabove and as shown in FIG. 3, the rising time becomes almost equal tothe falling time of the digital output signal V_(o1) and thedifferential output forces the pull-down transistor Q₃ to be driventemporarily and conductively. Therefore, the load driving capabilitybecomes very high in case the load capacitance CL₁ is light and even incase it is heavy, and the input impulse response can be improved becausethe load driving capability is high. Accordingly, since the rising timeof digital output signal V_(o1) is almost equal to the falling time,dependency on input waveform of digital input signal V_(i) for thecircuit speed of logic circuit LOG₁ can be reduced and the pull-downtransistor Q₃ equivalently forming the emitter load of emitter followeroutput circuit 2 is transitionally and conductively driven only when thedigital input signal V_(i1) changes to the high level V_(H) from the lowlevel V_(L) while the digital output signal V_(o1) changes to the lowlevel V_(L) from the high level V_(H), and in the steady state otherthan the transition period, the pull-down transistor Q₃ inhibits theemitter current according to keeping almost non-conductive state.Thereby, the logic circuit LOG₁ described above attains the high speedcharacteristic simultaneously with low power consumption.

In addition, in the logic circuit LOG₁ described, the emitter load inthe normal condition is equivalently in the high impedance condition andthereby a current flowing from the output load side is suppressed to alow level while the low level V_(L) output is obtained. Thereby, it canbe done easily to connect in common the outputs of a plurality of logiccircuits to form a wired logic circuit.

Moreover, in such logic circuit LOG₁, for example, even if the bipolartransistor Q₁ or Q₃ forming the phase split circuit 1 is effected by theα-ray, such influence of α-rays can be alleviated as explained for FIG.4.

FIG. 4 is a sectional view of a bipolar transistor Q₁ and collector loadresistance R₂ of logic circuit LOG₁ shown in FIG. 1.

In FIG. 4, P-Sub designates P type semiconductor substrate; N+--B1, N+type buried layer; N--E_(p) i, N type epitaxial layer; p--B1, baseregion of bipolar transistor Q₁ ; P--R, resistance layer of collectorload resistance R₂ ; N+--E1, emitter region of bipolar transistor Q₁ ;N+--C, collector contact layer of bipolar transistor Q₁ ; p+, channelstop region; D₄, depletion layer region and F, field oxide film. B, E, Crespectively designate aluminum electrodes of base, emitter andcollector of the bipolar transistor Q₁. R--A1 designates the electrodeof collector load resistance R₂ formed by aluminum and is connected tothe power source voltage V_(cc). The collector load resistance R₂ mayalso be formed by the polysilicon layer.

Under the steady condition, when the α-ray α is incident to the bipolartransistor Q₁, the holes ⊕ and electrons ⊖ are generated in thedepletion layer region D_(r). The electrons ⊖ are collected to thecollector potential, thereby providing the effect as it were region ofbipolar transistor Q₁ having the higher potential, thereby providing theeffect as it were applying a leak current to the P type semiconductorsubstrate from the collector region of the bipolar transistor Q₁. Inthis case, a virtual collector current generated by the α-ray lowers thecollector current of bipolar transistor Q₁ but almost does not have anyeffect on the emitter voltage of bipolar transistor Q₁ since the basevoltage is constant. Accordingly, the pull-down transistor Q₃ maintainsthe non-conductive condition without receiving the effect of the α-ray.While the pull-down transistor Q₃ sustains the non-conductive condition,the charges of load capacitance CL₁ are not discharged quickly even ifonly the collector voltage of bipolar transistor Q₁ is loweredtemporarily by the α-ray. Moreover, when the α-ray is incident to thepull-down transistor Q₃, the collector voltage of pull-down transistorQ₃ is caused to become low as in the case of explanation about operationof pull-down transistor Q₃ for the α-ray, but the collector of pull-downtransistor Q₃ is clamped by operations of emitter follower transistor Q₂and it does not drop largely. Therefore, the pull-down transistor Q₃maintains the non-conductive condition and the charges of loadcapacitance CL₁ is not discharged quickly. In addition, it is impossiblefor the α-ray entering the emitter follower transistor Q₁ (or pull-downtransistor Q₃) to be incident to the pull-down transistor Q₃ (or bipolartransistor Q₁) from the characteristic of the α-ray and it is almost farfrom possibility that a couple of α-rays are respectively incident tothe bipolar transistor Q₁ and pull-down transistor Q₃. Accordingly, thedigital output signal V_(o1) does not become low so much as the drop ofthe collector voltage of bipolar transistor Q₁, for example, even whenthe α-ray is incident to the bipolar transistor Q₁ or pull-downtransistor Q₃. Namely, the influence of the α-ray can be eased.

FIG. 5 shows a first embodiment for realizing the logic circuit LOG₁shown in FIG. 1 with the present invention. FIG. 6 illustrates thecharacteristic graph for explaining the relationship between the basevoltage V_(B3) and collector current I_(C3) of the pull-down transistorQ₃ included to the logic circuit LOG₁ shown in FIG. 5.

Explanation will be made focusing on the difference from FIG. 1. In thelogic circuit LOG₁ of an embodiment shown in the same figure, althoughnot limited particularly, the collector load resistance R₂, emitter loadresistance R₃ and resistance R₁ are respectively set to 3 kΩ, 2 kΩ and40 kΩ and the differential capacitance C_(al) is set to 0.2 PF.Moreover, in this embodiment, a high resistance R₄ of 20 kΩ is providedfor stabilizing the emitter voltage of the emitter follower transistorQ₂ in the steady state. Moreover, a bias circuit 4 which biases thepull-down transistor Q₃ in the steady state to the condition just beforethe conductive condition is also provided. This bias circuit 4 is formedby the bipolar transistor Q₄ to which the predetermined base controlvoltage V_(b1) supplied from the external or internal circuit of thecell formed by the logic circuit LOG₁ is applied and the predeterminedbias voltage V_(b) is supplied to the base of pull-down transistor Q₃.

Here, the predetermined base control voltage V_(b1) to be supplied tothe base of bipolar transistor Q₄ is set to such a predetermined voltageas will provide a voltage value for setting the pull-down transistor Q₃to the very weak ON condition, namely the bias voltage V_(b) which isgenerated the collector current I_(c3) of the pull-down transistor Q₃ toI_(c3') (for example, 10-100 μA) as shown in FIG. 6.

Thereby, if the driving sensitivity of pull-down transistor Q₃ isenhanced and the capacitance value of the differential capacitanceC_(al) is lowered to a small value, the pull-down transistor Q₃ isreliably driven conductively and the load capacitance CL₁ can be quicklydischarged at the transition time where the digital input signal V_(o1)is changed to the high level V_(H) from the low level V_(L).

As described, the falling time of the digital output signal V_(o1) tothe low level V_(L) from the high level v_(H) is further shortened andmoreover the load capacitance CL₁ can be discharged more quickly even incase the load capacitance CL₁ is a heavy load. Moreover, a conductivedriving level of the pull-down transistor Q₃ is determined by acapacitance value of differential capacitance C_(al) and the voltagevalue of bias voltage V_(b). Namely, the load driving capability of thelogic circuit LOG₁ can be controlled by changing the capacitance valueof differential capacitance C_(al) or the voltage value of base controlvoltage V_(b1) of bipolar transistor Q₄ and the falling rate of digitaloutput signal V_(o1) can also be controlled. The high impedanceresistance R₄ for stabilizing emitter voltage of emitter followertransistor Q₂ in the steady state is not always required in case thebase control voltage V_(b1) of the bipolar transistor Q₄ is set to aconstant value.

FIG. 7A illustrates waveforms indicating the simulation of input andoutput signals of the logic circuit LOG₁ shown in FIG. 5. FIG. 7B alsoillustrates waveforms indicating the collector current of emitterfollower transistor Q₂ and the collector current of pull-down transistorQ₃ during the simulation.

In such simulation, the collector load resistance R₂ is set to 1.9 kΩ,emitter load resistance R₃ to 1.3 kΩ, resistance R₁ to 20 kΩ,differential capacitance C_(al) to 0.7 pF, load capacitance CL₁ to 3 PF,base control voltage V_(b1) to -0.52 V, power source voltage V_(cc) to 0V and power source voltage V_(ee) to -1.985 V. The high resistance R₄shown in FIG. 5 is not provided for this simulation.

In FIG. 7A, the time is plotted on the horizontal axis in the unit ofseconds. On the other hand, the voltage of digital input signal V_(i1)and digital output signal V_(o1) is plotted on the vertical axis in theunit of volts. The digital input voltage V_(i1) and digital outputsignal V_(o1) respectively include some errors as shown in the waveformsof FIG. 7A, and the high level or low level including such errors arealso respectively indicated as the high level V_(H) and low level V_(L)in FIG. 7A. In FIG. 7B, the horizontal axis indicates the time like FIG.7A in the unit of seconds and the current of vertical axis indicates thecollector current I_(c2) of emitter follower transistor Q₂ and collectorcurrent I_(c3) of pull-down transistor Q₃ in the unit of amperes.

In FIG. 7A and FIG. 7B, when the digital input signal V_(i1) changes tothe high level V_(H) from the low level V_(L), the collector currentI_(c2) quickly increases due to the emitter follower operation of theemitter follower transistor Q₂ and thereby the load capacitance CL₁ isquickly charged, followed by the digital output signal V_(o1) rising upimmediately to the high level V_(H).

When the digital input signal V_(i1) changes to the high level V_(H)from the low level V_(L), the pull-down transistor Q₃ is temporarilydriven conductively with an output of the differential capacitanceC_(al) as described previously and thereby the collector current I_(c3)of pull-down transistor Q₃ quickly increases and the load capacitanceCL₁ is quickly discharged. Therefore, the digital output signal V_(o1)falls to the low level V_(L) from the high level V_(H) with a delay timet_(pd1') of about 0.6×10⁻⁹ sec.

FIG. 8 indicates an embodiment providing a plurality of differentialcapacitances in the logic circuit LOG₁ shown in FIG. 5. FIG. 9illustrates a layout of a plurality of differential capacitances shownin FIG. 8.

Explanation will now be made hereunder focusing on the difference fromthe embodiment of FIG. 5 with reference to FIG. 8. In the logic circuitLOG₁ shown in the same figure, the differential capacitances C_(alA),C_(alB) and C_(alC) are provided in place of the differentialcapacitance C_(al).

In FIG. 9, the regions Po and Po' enclosed by a broken line indicate thepolysilicon layer, respectively. A_(lA) -A_(lC) indicate wirings formedby the aluminum layer and C_(oA) -C_(oC) indicate contact holes.

The differential capacitances C_(lA), C_(lB) and C_(lC) are respectivelyformed in the doubled polysilicon layers formed from the polysiliconlayers Po, Po', interposing dielectric films not illustrated. Thewirings A_(lA) -A_(lC) and contact holes C_(oA) -C_(oG) coupled with thedifferential capacitances C_(alA), C_(alB) and C_(alC) are selected byCAD (Computer Aided Design) and DA (Design Automation). The differentialcapacitances C_(alA), C_(alB) or C_(alC) are coupled between the emitterof bipolar transistor Q₁ and the base of pull-down transistor Q₃ withthe mask to form the pattern of selected wirings A_(lA), A_(lB) orA_(lC) and contact holes C_(oA), C_(oB) or C_(oG).

As described above, the logic circuit LOG₁ can select adequate loaddriving capability and falling rate of the digital output signal V_(o1)for the load capacitance CL₁ by selecting the differential capacitancesC_(alA), C_(alB) or C_(alC).

FIG. 10 shows an embodiment indicating an application example of thelogic circuit LOG₁ shown in FIG. 5 to which the present invention isapplied.

Focusing to difference from FIG. 5, in the logic circuit LOG₁, thebipolar transistor Q₁ forming the phase split circuit 1 comprises thebipolar transistors Q_(lA) and Q_(lB) forming a 2-input logic circuitand the digital input signals V_(i1A) and V_(i1B) are respectivelysupplied to the bases of bipolar transistors Q_(lA) and Q_(lB).

Operations of embodiment shown in FIG. 10 will then be explainedhereunder.

When the digital input Signals V_(i1A) and V_(i1B) are all set to thelow level V_(L), the voltage of emitters connected in common of thebipolar transistors Q_(lA) and Q_(lB) becomes the low level and acomparatively small collector current flows into the collector loadresistance R₂. Therefore, the node n₁ becomes high level V_(HC) like theembodiment of FIG. 1, while the digital output signal V_(o1) becomespredetermined high level V_(H).

On the other hand, any one of the digital input signals V_(i1A) andV_(i1B) becomes high level V_(H), the voltage of emitters connected incommon of the transistors Q_(lA) and Q_(lB) becomes high level and acomparatively large collector current flows into the collector loadresistance R₂. Thereby, the digital output signal V_(o1) is set to thepredetermined low level V_(L).

Namely, the logic circuit LOG₁ of this embodiment functions as the2-input NOR gate circuit in which the digital output signal V_(o1)satisfies following logical expression. ##EQU1##

As described above, in the embodiment of FIG. 5, the bipolar transistorQ₁, of the logic circuit LOG₁, forming the phase split circuit 1 isprovided in parallel and these are replaced by a couple of transistorsQ_(lA) and Q_(lB) which respectively receive the digital input signalsV_(i1A) and V_(i1B) corresponding to the bases of these transistors.When the digital input signals V_(i1A) and V_(i1B) are all low levelV_(L), the digital output signal V_(o1) is set selectively to the highlevel V_(H) and thereby the logic circuit LOG₁ of this embodimentfunctions as the 2-input NOR gate. It is natural that the logic circuitLOG₁ realizes low power consumption, like the embodiment of FIG. 1,without interference on the high operation rate. In addition, withincrease or decrease in the number of bipolar transistors provided inparallel, the NOR gate circuit having a desired fan-in number can berealized.

FIG. 11 illustrates a first application example comprising a wired logic5 by connecting a couple of 2-input logic circuits LOG₁ show in FIG. 10,as described, as the cell of large capacity gate array integratedcircuit. In FIG. 11, the resistance R₄ shown in FIG. 10 may also beprovided separately, although not illustrated. Moreover, since a pair oflogic circuits LOG₁ are indicated, the one is designated as LOG_(1A) andthe other as LOG_(1B). Moreover, the digital input signals V_(i1A),V_(i1B) are also indicated in the manner that the digital input signalsof logic circuit LOG_(1A) as V_(i1A), V_(i1B) and the digital inputsignals of logic LOG_(lB) as V_(i1A'), V_(i1B'), corresponding to thelogic circuits LOG_(lA), LOG_(lB).

As is already described, the emitter load is equivalently in the highimpedance during the steady state and a current flowing from the loadside during output of the low level V_(L) is suppressed to a smalllevel. Thereby, it can be realized easily to form a wired logic 5 byconnecting in common the respective outputs without flowing of a steadycurrent to the wires l_(i) and l₂. Namely, the digital output signalV_(o2) indicated below can be realized by forming the wired logics.

    V.sub.o2 =(V.sub.i1A +V.sub.i1B +V.sub.i1A' +V.sub.i1B')

Moreover, if the wires 11 and 12 are formed by aluminum, for example,the electromigration is not easily generated because a steady currentdoes not flow and thereby superminiaturization of wires l₁ and l₂ can berealized and in addition high speed operation and high integrationdensity of large capacity gate array integrated circuit can also berealized.

The electromigration is explained, for example, on page 393, of the bookentitled "SEMICONDUCTOR DEVICE--BASIC THEORY AND PROCESS TECHNOLOGY",issued on May 25, 1987, SANGYO TOSHO Inc.

FIG. 12 is a second application example comprising a latch circuit usinga pair of 2-input logic circuit LOG₁ shown in FIG. 10 as the cell of thelarge capacity gate array integrated circuit as described previously.

In FIG. 12, the resistance R₄ shown in FIG. 10 may also be providedseparately, although not illustrated. Moreover, a pair of logic circuitsLOG₁ are provided in FIG. 12 and the one is designated as LOG_(1c),while the other as LOG_(1D).

As is already described, both logic circuits LOG_(1C), LOG_(1D) assurethe storing Operations with very high reliability because the digitaloutput signal V_(o1) does not easily receive the influence of α-ray.

FIG. 13 indicates an embodiment in which the logic circuit LOG₁ shown inFIG. 10 and the logic circuit LOG₂ forming the ECL are respectivelycombined as the cells of large capacity gate array integrated circuit.

In FIG. 13, the logic circuit LOG₂ is formed by differential transistorsQ₅ and Q₆ coupled between the power source voltage V_(cc) and powersource voltage V_(eel') differential amplifier circuit 6 formed byresistances R₅, R₆ and constant current source IS₁ and emitter followeroutput circuit 7 formed by bipolar transistors Q₇ and Q₈ which arecoupled between the power source voltage V_(cc) and power source voltageV_(ee) and receive output of the differential amplifier circuit 6 andresistances R₇ and R₈. Moreover, the digital output signal V_(o1) oflogic circuit LOG₁ is supplied to the base of bipolar transistor Q₅ andthe reference voltage V_(refl) to the base Of bipolar transistor Q₆. Inthe logic circuit LOG₁, the high resistance R₄ of FIG. 10, although notillustrated, may be provided by different manner. The power sourcevoltage V_(eel) is set to the predetermined negative voltage and forexample set to -3 V.

Operations of the logic circuits LOG₁, LOG₂ in the embodiment of FIG. 13are explained hereunder.

The logic circuit LOG₁ is the same as the embodiment shown in FIG. 10and therefore it is omitted here. When the digital output signal V_(o1)of logic circuit LOG₁ is in the high level V_(H), it conductively drivesthe bipolar transistor Q₅ because it is higher than the potential ofreference voltage V_(refl) and the digital output signals V_(o3), V_(o4)of the emitter follower output circuit 7 respective become the low levelV_(L) and high level V_(H). When the digital output signal V_(o1) oflogic circuit LOG₁ is low level V_(L), it conductively drives thebipolar transistor Q₆ because it is lower than the reference voltageV_(refl) and the digital output signals V_(o3), V_(o4) of the emitterfollower output circuit are respectively output as the high level V_(H)and low level V_(L).

The logic circuit LOG₁ and the logic circuit LOG₂ forming ECL can bearranged simultaneously on a semiconductor substrate of the largecapacity gate array integrated circuit by adequately providing theresistances R₁ -R₈ in Order to assure compatibility of the logicamplitudes of the high level and low level of the digital output signalV_(o1) of the logic circuit LOG₁ and the logic amplitude of the logiccircuit LOG₂.

FIG. 14 indicates an embodiment for stabilizing outputs of logic circuitLOG₁ shown in FIG. 10. FIG. 15 is a layout of the logic circuit LOG₁shown in FIG. 14.

In FIG. 14, explanation is made focusing on the difference from FIG. 10.The logic circuit LOG₁ comprises a bias circuit 4 for biasing thepull-down transistor Q₃ in the steady state to the condition just beforethe conductive condition and a clamp circuit 8 to prevent overshoot ofthe digital output signal V_(o1) to the negative voltage side which maybe generated when the pull-down transistor temporarily becomesconductive and the discharge current at this time is extractedexcessively. Moreover, a pair of power supply lines are respectivelyprovided for the power source voltages V_(cc) and V_(ee) in Order toprevent malfunction by conductivity at the steady state of the pull-downtransistor Q₃ due to variation of power source voltages V_(ee) andV_(cc) depending on temperature characteristic of the power supply linesof power source voltages V_(ee) and V_(cc), and the power sourcevoltages supplied from respective power supply lines are indicated asV_(cc1), V_(cc2), V_(ee2) and V_(ee3).

The clamp Circuit 8 is formed by the bipolar transistor Q₉ to which thepredetermined base control voltage V_(b) of low impedance is appliedfrom the logic circuit not illustrated, and the voltage of digitaloutput signal V_(o1) is boosted above the low level V_(L) with theemitter follower operation of transistor Q₉.

In FIG. 15, N+--B1 of the region enclosed by the dotted line designatesN+ type burred layer; N+--E2, the emitter region of correspondingbipolar transistors. The region P--B2 enclosed by a broken linedesignates the base region of corresponding bipolar transistor; theregions Po, Po' enclosed by the broken line respectively designate thepolysilicon layer, the region enclosed by a solid line designates thewirings Al₁ -Al₁₂ formed by aluminum and each Co, the contact B₁ -B₄ andB₉, E₁ -E₄ and E₉ designate the base and emitter of the bipolartransistors Q₁ -Q₄ and Q₉ ; C₃ is collector of the emitter followertransistor Q₃ ; CC₁ and CC₂ designate the common collectors describedlater. The collector load resistance R₂ is not a resistance formed bythe diffused layer shown in FIG. 4 but the resistance formed by thepolysilicon layer.

In FIG. 15, the digital input Signals V_(i1A) and V_(i1B) are suppliedto the bases B_(1A) and B_(1B) of bipolar transistors Q_(1A), Q_(1B)through the corresponding wirings Ali and Al₂ formed by aluminum layer.The power source voltage V_(ee2) is coupled to the emitters E_(1A) andE_(1B) of the bipolar transistors Q_(1A), Q_(1B) through the wirings Al₃and Al₄ formed by aluminum layer and the load resistance R₃ formed bythe polysilicon layer P_(o). Moreover, these emitters El_(A), El_(B) arealso coupled to the one electrode of differential capacitance C_(al)through the wiring Al₄ formed by the aluminum layer. The collectors ofbipolar transistors Q_(1A), Q_(1B) are formed as the common collectorCC₁ and is connected to the power source voltage V_(cci) through thewirings Al₅ and Al₆ and collector load resistance R₂ formed by thepolysilicon layer P_(o). This common collector CC₁ is coupled to thebase B₂ of emitter follower transistor Q₂ through the wiring Al₆ formedby the aluminum layer.

The differential capacitance C_(al) is formed by interposing adielectric film not illustrated between the doubled polysilicon layersformed by the polysilicon layers P_(o), P_(o) ' and the one electrode ofthe differential capacitance C_(al) is coupled to the emitters E_(1A)and E_(1B) of the bipolar transistors Q_(1A), Q_(1B) and emitter loadresistance R₃ through the wiring Al₄ formed by the aluminum layer. Theother electrode of differential capacitance C_(al) is coupled to thebase B₃ of pull-down transistor Q₃, resistance R₁ formed by polysiliconlayer P_(o) and emitter E₄ of bipolar transistor Q₄ through the wiringAl₇ formed by the aluminum layer.

The pull-down transistor Q₃ is connected, at its base B₃, to thedifferential capacitance C_(al), resistance R₁ and emitter E₄ of bipolartransistor Q₄ through the wiring Al₇ formed by the aluminum layer, asdescribed above. The emitter E₃ of pull-down transistor Q₃ is connectedto the power source voltage V_(ee3) and the resistance R₁ formed by thepolysilicon layer P_(o) through the wiring Al₈ formed by the aluminumlayer. The collector C₃ of pull-down transistor Q₃ is connected to theemitter E₂ of emitter follower transistor Q₂, emitter E₉ of bipolartransistor Q₉ and the resistance R₄ formed by the polysilicon layerP_(o) through the wiring Al₉ formed by the aluminum layer and thiswiring Al₉ of aluminum layer is used for outputting the digital outputsignal V_(o1).

The emitter E₂ of emitter follower transistor Q₂ is connected, asdescribed above, to the collector C₃ of pull-down transistor Q₃,resistance R₄, and emitter E₉ of bipolar transistor Q₉. The base B₂ ofemitter follower transistor Q₂ is connected, as described, to the commoncollector CC₁ and collector load resistance R₂ of the bipolartransistors Q_(1A), Q_(1B) through the wiring Al₆ formed by thealuminum. The collector of emitter follower transistor Q₂ is coupled incommon with the collectors CC₂ of bipolar transistors Q₄, Q₉ and thiscommon collector is connected to the power source voltage V_(cc2)through the wiring Al₁₀ formed by the aluminum layer.

Regarding the emitter E₄ and collector of the bipolar transistor Q₄, theemitter E₄ is coupled to the base B₃ of pull-down transistor Q₃,differential capacitance C_(al) and resistance R₁ through the wiring Al₇formed by the aluminum layer, while the collector is formed as thecommon collector CC₂ together with the collectors of emitter followertransistor Q₂ and bipolar transistor Q₄. The base B₄ of bipolartransistor Q₄ is connected to the wiring Al₁₁ formed by the aluminumlayer and the base control voltage Vb₁ is supplied through this wiring.

The emitter E₉ and collector of the bipolar transistor Q₉ are connected,as described above, to the resistance R₄, emitter E₂ of emitter followertransistor Q₂ and collector C₃ of pull-down transistor Q₃ through thewiring Al₉ formed by the aluminum layer, while the collector is formedas the common collector CC₂ together with the collector of bipolartransistor Q₉ and collector of emitter follower transistor Q₄. The baseB₉ of bipolar transistor Q₉ is coupled to the wiring Al₁₂ formed by thealuminum layer and the base control voltage V_(b2) is supplied throughthis wiring.

As described above, the overshoot of digital output signal V_(o1) towardthe negative voltage is prevented and low level V_(L) of digital outputsignal V_(o1) can be ensured. Moreover, since the collectors of bipolartransistors Q_(1A) and Q_(1B) forming the phase split circuit 1 or thecollectors of bipolar transistor Q₄ of bias circuit 4 and the bipolartransistor Q₉ of clamp circuit 8 are formed on the same region of asemiconductor substrate, thereby realizing a superminiaturized cell.

FIG. 16 indicates a second embodiment for realizing the first principlediagram of the logic circuit LOG₁ shown in FIG. 1 to which the presentinvention is applied.

FIG. 16 will be explained focusing on difference from the embodiment ofFIG. 1. The logic circuit LOG₁ of this embodiment also is provided witha bias circuit 9.

The bias circuit 9 is formed by the resistance R₉ coupled between theemitter of emitter follower transistor Q₂ and the differentialcapacitance C_(al). This bias circuit 9 increases a charging current fordriving the pull-down transistor Q₃ with the differential capacitanceC_(al) by DC-feedback of the digital output signal V_(o1) to the base ofpull-down transistor Q₃ through the resistance R₉ when the digitaloutput signal is in the high level V_(H).

Thereby, good falling characteristic can be obtained even when the loadcapacitance CL₁ changes.

FIG. 17 indicates an embodiment of the wired logic 10 formed by wiring acouple of logic circuits LOG₁ shown in FIG. 16 as the cells of largecapacity gate array integrated circuit.

In FIG. 17, Since a pair of logic circuits LOG₁ are shown, the one isdesignated as LOG_(1E) and the other as LOG_(1F). Moreover, the digitalinput signal V_(i1) is also indicated as V_(i1) and V_(i1'),respectively for the logic circuits LOG_(1E) and LOG_(1F). As is alreadydescribed, a pair of logic circuits LOG_(1E), LOG_(1F) can easily formthe wired logic 10 since a current flowing from the load side when theoutput is low level V_(L) is suppressed to a lower level.

In FIG. 17, the digital output signal V_(o5) =(V_(i1) +V_(i1')) for thedigital input signals V_(i1), V_(i1'), can be realized by forming thewired logic 10. In the same figure, the wired logic 10 is formed bynon-connection of a part of the cell formed by the logic circuitLOG_(1F) namely non-application of the bias circuit 9, resistance R₁ andpull-down transistor Q₃ of the logic circuit LOG_(1D). Accordingly, acurrent flowing from the load side is suppressed smaller than that inthe embodiment of FIG. 11 and moreover electromigration of wirings l₃and l₄ is not easily generated and superminiaturization can then berealized.

FIG. 18 indicates a third embodiment for realizing the first principlediagram of the logic circuit LOG₁ shown in FIG. 1 by the presentinvention.

Explanation will be made focusing on difference from FIG. 1. The logiccircuit LOG₁ is formed as a load means of the emitter followertransistor Q₂ forming the output emitter follower circuit and moreoverwith addition of serial resistances R₁₀ and R₁₁. A resistance R₁₂ isprovided, although not particularly limited, between the node connectedin common of the resistances R₁₀ and R₁₁ and the base of pull-downtransistor Q₃. The base of pull-down transistor Q₃ is coupled withcapacitance to the emitter of the bipolar transistor Q₁ through thedifferential capacitance C_(al). Here, the resistances R₁₀ and R₁₁ aredesigned to have comparatively large resistance values and theresistance R₁₂ is designed to have a predetermined resistance valuewhich makes sufficiently large an input impedance of the pull-downtransistor Q₃. Thereby, the resistance R₁₂ forms a bias circuit 11,together with the resistances R₁₀ and R₁₁, which gives the predeterminedbias voltage V_(b), to the base of pull-down transistor Q₃ and alsoforms a differential circuit, together with the differential capacitanceC_(al), which transmits level change of digital input signal V_(i1) tothe base of pull-down transistor Q₃.

When the digital input signal V_(i1) is fixed to the predetermined lowlevel V_(L) and the digital output signal V_(o1) to the predeterminedhigh level V_(H), the bias voltage V_(b), predetermined by theresistance ratio of resistances R₁₀ and R₁₁ is applied to the base ofpull-down transistor Q₃. This bias voltage V_(b), is set to thepredetermined voltage value which makes the pull-down transistor Q₃ toweak ON condition as in the case of the embodiment shown in FIG. 5.Thereby, the pull-down transistor Q₃ is set to the very weak ONcondition without giving influence on the digital output signal V_(o1).

When the digital input signal V_(i1) changes to the high level V_(H)from the low level V_(L), the base voltage of pull-down transistor Q₃becomes temporarily high since level change of of digital input signalV_(i1) is transmitted through the differential circuit formed by thedifferential capacitance C_(al) and resistance R₁₂. Therefore, thepull-down transistor Q₃ becomes perfect ON condition temporarily and thedigital output signal V_(o1) is quickly Changed to the low level V_(L)from the high level V_(H).

While the digital input signal V_(i1) is fixed to the predetermined highlevel V_(H) and the digital output signal V_(o1) to the predeterminedlow level V_(L), the base voltage of pull-down transistor Q₃ is set tothe low level and the pull-down transistor Q₃ is set to almost thecut-off condition. Under this condition, when the digital input signalV_(i1) changes to the low level V_(L) from the high level V_(H), thebase voltage of pull-down transistor Q₃ is temporarily lowered becauselevel change of digital input signal V_(i1) is transmitted through thedifferential circuit. Thereby, the transistor Q₃ is further set to deepcut-off condition and the digital output signal V_(o1) is quicklychanged to the high level from the low level V_(L) through the emitterfollower transistor Q₂.

As described above, the base of pull-down transistor Q₃ in the logiccircuit LOG₁ of this embodiment is capacitively coupled to the emitterof bipolar transistor Q₁ through the differential capacitance C_(al) andalso coupled to the node connected in common of the resistances R₁₀ andR₁₁ through the resistance R₁₂. The resistance R₁₂ forms thedifferential circuit, together with the differential capacitance C_(al),which sends level change of digital input signal V_(i1) to the base ofpull-down transistor Q₃ and also forms a bias circuit 11, together withthe resistances R₁₀ and R₁₁, which gives the predetermined bias voltageV_(b), to the base of pull-down transistor Q₃. The logic circuit LOG₁ ofthis embodiment enhances the driving sensitivity of the pull-downtransistor Q₃ by supplying the bias voltage V_(b), to the base ofpull-down transistor Q₃. The malfunction due to the change of powersource voltage V_(ee) of pull-down transistor Q₃ shown in FIG. 14 can bereduced by supplying the bias voltage V_(b), with the digital outputsignal V_(o1).

FIG. 19 indicates an embodiment in which a constant voltage circuit isadded to the logic circuit LOG₁ shown in FIG. 5.

Emphasis is placed on difference from FIG. 5 for the followingexplanation. In the same figure, a diode D₁ is provided between the baseof bipolar transistor Q₄ and power source voltage V_(cc). Moreover, aresistance R₁₃ is provided, although not particularly limited, betweenthe base of bipolar transistor Q₄ and the power source voltage V_(ee),while a capacitance C_(a2) between the base of bipolar transistor Q₄ andthe emitter of emitter follower transistor Q₂, namely the outputterminal of the logic circuit LOG₁. Thereby, the resistance R₁₃ forms,together with the diode D₁, a constant voltage generating circuit whichgives the predetermined base control voltage V_(b3) to the base ofbipolar transistor Q₄ and also forms, together with the capacitanceC_(a2), a differential circuit which sends level change at the outputterminal to the base of bipolar transistor Q₄. Here, the resistance R₁₃is designed to have a comparatively large resistance value and does notprevent power saving of the logic circuit LOG₁.

When the level Of digital output signal V_(i1) is fixed to the highlevel V_(H) or low level V_(L), a predetermined base control voltageV_(DF1) which is determined by the forward voltage V_(DF1) of diode D₁is supplied to the base of bipolar transistor Q₄. This base controlvoltage V_(b3) is set, as in the case of the base control voltage V_(b1)in the embodiment shown in FIG. 5, to the predetermined voltage valuewhich applies a bias voltage V_(b) to the base of pull-down transistorQ₃ so that it is set to a very weak ON condition. From these facts, thepull-down transistor Q₃ functions similar to the embodiment of FIG. 5and thereby the logic circuit LOG₁ of this embodiment realizes low powerconsumption without giving influence on the high speed operationcharacteristic.

Here, a differential circuit consisting of the capacitance C_(a2) andresistance R₁₃ is provided, as described above, between the outputterminal of logic circuit LOG₁ of this embodiment and the base ofbipolar transistor Q₄. This differential circuit sets the digital outputsignal V_(o1) to the low level V_(L), causes discharging of differentialcapacitance C_(a1) and improves response characteristic for thepulsewise input signal.

As described, in the logic circuit LOG₁ of this embodiment, the diode D₁is provided between the base of bipolar transistor Q₄ forming the biascircuit 4 and the power source voltage V_(cc) and a capacitor C_(a2) isprovided between the base thereof and the output terminal of the logiccircuit LOG₁. Moreover, the resistance R₁₃ which forms a constantvoltage generating circuit together with the diode D₁ and forms adifferential circuit together with the capacitance C_(a2) is providedbetween the base of bipolar transistor Q₄ and the power source voltageV_(ee).

This resistance R₁₃ is designed to have a comparatively large resistancevalue. Thereby, the logic circuit LOG₁ of this embodiment simplifies theconstant voltage generating circuit for giving the predetermined basecontrol voltage V_(b3) to the base of bipolar transistor Q₄ and furtherimproves response characteristic for the pulsewise input signal byfeedback operation to be carried out through the bipolar transistors Q₄and Q₃. As described above, since the resistances R₄ and R₁₃ aredesigned to have comparatively large resistance value, low powerconsumption of the logic circuit LOG₁ can be realized.

FIG. 20 is the second principle diagram of the logic circuit to whichthe present invention is applied.

In FIG. 20, this circuit is basically the same as the embodiment of FIG.1 and the phase split circuit 1, emitter follower output circuit 2 andactive pull-down circuit 3, etc. correspond to those in the embodimentof FIG. 1.

Following explanation will be made placing emphasis on difference fromFIG. 1. In the same figure, the phase split circuit 1 is formed by thebipolar transistors Q_(1c) and Q_(1D) which receive the digital inputsignals V_(i1C) and V_(i1D). Moreover, even in case the logic of digitalinput signals V_(i1C), V_(i1D) changes to the high level V_(H) and lowlevel V_(L) with a little time difference, the digital output signalV_(o1) quickly responds to such change. Therefore, a capacitance C_(a3)is provided for urging discharge operation of differential capacitanceC_(al) for conductively driving the emitter load transistor Q₃.

FIG. 21 illustrates the waveforms for explaining the operations of logiccircuit LOG₁ shown in FIG. 20.

In the operation example of the same figure, since the logic of a pairof digital input signals V_(i1C), and V_(i1D) changes to the high levelV_(H) only with a little time difference, the digital output signalV_(o1) becomes high level V_(H) only within a very short period.

In FIG. 20 and FIG. 21, when, in the period to wherein the digital inputsignal V_(i1c) and V_(i1d) are respectively high level V_(H) and lowlevel V_(L), the differential capacitance C_(al) is charged to thesaturated condition, viewed from the side of phase split circuit 1,because the common emitter voltage of bipolar transistors Q_(1c),Q_(1D), namely the non inverted output +V_(i) of phase split circuit 1is high level. Moreover, the capacitance C_(a3) is in the nonchargedcondition viewed from the output side because the emitter voltage ofemitter follower transistor Q₂, namely the digital output signal V_(o1)is low level.

Next, in the period t₁ wherein the digital input signal V_(i1C) changesto the low level V_(L) from the high level V_(H) and the digital inputsignals V_(i1c) and V_(i1D) become low level V_(L), the non-invertedoutput +V_(i) of the phase split circuit 1 falls to the low level fromthe high level and thereby, the differential capacitance C_(al) in thecharging condition starts in turn the discharge operation.Simultaneously, since the emitter voltage of emitter follower transistorQ₂, namely the digital output signal V_(o1) rises to the high levelV_(H) from the low level V_(L), the capacitance C_(a3) in thenon-charging condition starts in turn the charging operation. That is,discharge operation of the differential capacitance C_(al) isaccelerated by the charging Operation of capacitance C_(a3). As aresult, the differential capacitance C_(al) charged is sufficientlydischarged within a very short period from the timing where thenon-inverted output +V_(i) of the phase split circuit 1 falls to the lowlevel from the high level.

Thereby, in the period t₂ where the digital input signal V_(i1D) changesto the high level V_(H) from the low level V_(L) and the non-invertedoutput +V_(i) of phase split circuit 1 rises to the high level from thelow level, the differential capacitance C_(al) allows the chargingcurrent to flow until sufficiently driving the pull-down transistor Q₃.Accordingly, the digital output signal V_(o1) can immediately be loweredto the low level from the high level.

Here, if the capacitance C_(a3) is not provided, since the period t₁ inwhich the logical OR of the digital input signals V_(i1C) and V_(i1D)becomes the low level is very short period, the differential capacitanceC_(al) cannot sufficiently discharge the electric charge charged whenthe non-inverted output +V_(i) is high level. Therefore, in the periodin which the non-inverted output +V_(i) changes to the high level fromthe low level, namely in the period t₂ in which the digital outputsignal V_(o1) changes to the low level from the high level, thedifferential capacitance C_(al) can no longer supply the sufficientcharging current to the base of pull-down transistor Q₃. As a result, asindicated by a dotted line in FIG. 21, level shift to the low level ofthe digital output signal V_(o1) is much delayed.

As described above, even when the digital input signals V_(i1C) andV_(i1D) of the logic circuit LOG₁ of the embodiment shown in FIG. 20change within a very short period, such level change can be sentcertainly to the digital output signal V_(o1).

FIG. 22 indicates an embodiment for realizing the logic circuit LOG₁shown in FIG. 20 by the present invention. Focusing on the differencefrom FIG. 20, in the logic circuit LOG₁ of the embodiment shown in thesame figure, a clamp circuit 13 which rejects overshoot of digitaloutput signal V_(o1) in the negative voltage side like the clamp circuit8 shown in FIG. 14 and a resistance R₁₄ for assuring the level ofdigital output signal V_(o1) like the high resistance R₄ shown in FIG. 5are provided, together with the bias circuit 12 which biases thepull-down transistor Q₃ in the steady state to the condition immediatelybefore the conductive condition like the bias circuit shown in FIG. 5.Like the circuit of FIG. 5, the bias circuit 12 is configurated by thebipolar transistor Q₁₀ to which the predetermined base control voltageV_(b4) supplied from the external or internal circuit of the cell formedby the logic circuit LOG₁ and the predetermined bias voltage V_(b'), issupplied to the base of pull-down transistor Q₃ by applying thepredetermined bias current to the resistance R₁ connected in parallel tothe base of pull-down transistor Q₃. Like the bias voltage V_(b) shownin FIG. 5 and FIG. 6, the predetermined bias voltage V_(b'), is avoltage value for setting the pull-down transistor Q₃ to the very weakON condition, namely a voltage value which result sin the collectorcurrent I_(c3) of the pull-down transistor of about 10-100 μA.

The clamp circuit 13 is formed by the bipolar transistor Q₁₁ to whichthe predetermined base control voltage V_(b5) is applied like thecircuit shown in FIG. 14 and the voltage of digital output signal V_(o1)is boosted exceeding the predetermined low level V_(L) by the emitterfollower operation of the bipolar transistor Q₁₁.

FIG. 23 indicates an embodiment of the second logic circuit to which thepresent invention is applied.

In FIG. 23, the logic circuit LOG₃ is formed by the phase invertingcircuit 14 which receives the complementary digital input signalsV_(i2), V_(i3) and outputs the phase-inverted digital input signalsV_(i2), V_(i3), emitter follower output circuits 15, 16 which are drivenby outputs of the phase inverting circuit 14, active pull-down circuits17, 18, differential capacitances C_(a4), C_(a5) for differentiatingrising edges of the digital input Signals V_(i2), V_(i3) suppliedthrough bipolar transistors Q₁₅, Q₁₆, and bias circuits 19, 20.

The phase inverting circuit 14 is formed by the bipolar transistors Q₁₂-Q₁₄ and resistances R₁₅ -R₁₇, the predetermined base control voltageV_(b6) (for example, 1.85 V) is supplied to the base of bipolartransistor Q₁₄ and thereby it forms a constant current circuit togetherwith the resistance R₁₇. As described the phase inverting circuit 14inverts the phase of the complementary digital input signals V_(i2),V_(i3) and outputs such signals to the emitter follower output circuits15, 16.

The emitter follower output circuits 15, 16 are respectively formed bythe emitter follower transistors Q₁₇, Q₁₈. The emitters of emitterfollower transistors Q₁₇, Q₁₈ are respectively connected to the powersource voltage V_(ee) through the pull-down transistors Q₁₉, Q₂₀described later and also connected to the load capacitance CL₂ generatedequivalently by the distributed capacitance of the output wirings.

The active pull-down circuits 17, 18 are respectively formed by thepull-down transistors Q₁₉, Q₂₀. The pull-down transistors Q₁₉, Q₂₀ areemitter loads of the emitter output circuits 15, 16 and switchinglyconnect the output of emitter follower transistors Q₁₇, Q₁₈, namely thedigital output signals V_(o6), V_(o7) of the logic circuit LOG₃respectively to the power source voltage V_(ee)..

The differential capacitances C_(a4), C_(a5) differentiate the in-phasesignals of the digital input signals V_(i2), V_(i3) supplied through thebipolar transistors Q₁₅, Q₁₆. when the respective digital input signalsV_(i2), V_(i3) rise to the high level from the low level andconductively drive the pull-down transistors Q₁₉, Q₂₀ temporarily.

The bias circuits 19, 20 comprise the resistances R₂₂, R₂₃ which feedthe voltages of digital output signals V_(o6), V_(o7) back to the basesof pull-down transistors Q₁₉, Q₂₀ and the resistances R₂₀, R₂₁ whichpull down the bases of pull-down transistors Q₁₉, Q₂₀ to the side ofemitter.

The bipolar transistors Q₁₅, Q₁₆ form the emitter follower to therespective digital input signals V_(i2), V_(i3), buffer-amplify thedigital input signals V_(i2), V_(i3) in the in-phase condition and thensends such signals to the differential capacitances C_(a4), C_(a5).

As described previously, the power source voltages V_(ee), V_(ee1) arenegative power source voltages and set, for example, to -2 V, -3 V,respectively. In addition, the power source voltage V_(cc) is the groundpotential as described previously.

Operations of logic circuit LOG₃ thus configurated as described abovewill be explained hereunder.

In FIG. 23, when the digital input signal V_(i2) is low level, the baseinput of the emitter follower transistor Q₁₇ of the emitter followeroutput circuit 15 becomes high level. Thereby, the emitter followertransistor Q₁₇ is conductively driven and the digital output signalV_(o6) becomes high level. In this timing, the pull-down transistor Q₁₉keeps the non-conductive condition.

Here, when the digital input signal V_(i2) changes to the high levelfrom the low level, the base input of the emitter follower transistorQ₁₇ becomes the low level from the high level and the emitter followertransistor Q₁₇ is Changed to the non-conductive condition from theconductive condition. In this case, when the digital input signal V_(i2)rises to the high level from the low level, a charging current I₃ of thedifferential capacitance C_(a4) flows into the pull-down transistor Q₁₉.

Accordingly, the pull-down transistor Q₁₉ becomes conductive temporarilywhile the emitter follower transistor Q₁₇ changes to the conductivecondition from the non-conductive condition, forcing the charges of loadcapacitance C_(L2) to quickly discharge. Thereby, the digital outputsignal V_(o6) immediately falls to the low level from the high levelafter the digital input signal V_(i2) changes to the high level from thelow level.

In the same way, the digital output signal V_(o7) also falls immediatelyto the low level from the high level after the digital input signalV_(i3) changes to the high level from the low level.

As described previously, when the digital output signals V_(o6),V_(o7)fall to the low level from the high level, the pull-down transistorsQ₁₉, Q₂₀ are temporarily set to the conductive condition with thedifferential output. Thereby, the load capacitance CL₂ is forced todischarge and the digital output signals V_(o6), V_(o7) quickly fall. Inother cases, the pull down transistors Q₁₉, Q₂₀ are kept at the nonconductive condition and a current steadily consumed is kept small.

Further, in the logic circuit LOG₃, the base bias condition of pull-downtransistors Q₁₉, Q₂₀ changes depending on the load condition of thedigital output signals V_(o6), V_(o7). Namely, when a load capacitanceCL₂ is still charged with remaining charges, the conductive drivinglevel of the pull-down transistors Q₁₉, Q₂₀ is enhanced because the highvoltage by the remaining charges increases the base bias level of thepull-down transistors Q₁₉, Q₂₀. Thereby, the pull-down effect, namelythe effect of lowering the digital output signals V_(o6), V_(o7) canalso be enhanced.

In addition, when the digital output signals V_(o6), V_(o7) are highlevel, a charging current I₄ is supplied to charge the differentialcapacitances C_(a4), C_(a5) output signals V_(o6), V_(o7). Therefore,the charging current for driving the pull-down transistors Q₁₉, Q₂₀ fromthe differential capacitances C_(a4), C_(a5) also increases by DCfeedback of the voltage of digital output signals V_(o6), V_(o7) to thebase to pull-down transistors Q₁₉, Q₂₀.

Thereby, good falling characteristic of the logic circuit LOG₃comprising the emitter follower output circuits 15, 16 can be obtainedwithout relation to change of load capacitance CL₂.

FIG. 24 shows an embodiment indicating a third logic circuit to whichthe present invention is applied.

In FIG. 24, the emitter follower output circuit 15, active pull-downcircuit 17, bias circuit 19 and differential capacitance C_(a4)correspond to those of embodiment shown in FIG. 23 and operate in thesame way.

Moreover in FIG. 24, the logic circuit LOG₄ is formed by the D latchcircuit 21 which is configurated using the clocked gate by the 2-stagecascade differential circuit consisting of the bipolar transistors Q₂₁-Q₂₅ and resistances R₂₄ -R₂₆ and emitter follower output circuit formedby the bipolar transistors Q₂₈, Q₂₉ which provide outputs of D latchcircuit 14 as the latch outputs Q, Q, and the resistances R₂₇, R₂₈.

The D latch circuit 21 executes the D latch operations by the internalfeedback of outputs of clocked gate through the emitter follower by theemitter follower transistor Q₁₇. The bipolar transistor Q₂₅ forms aconstant current source, together With the resistance R₂₆, by receivingthe predetermined base control voltage V_(b7) (for example, -1.85 V).The digital input signal V_(i4) is data input; CK is clock input; +Q, -Qare latch outputs; V_(ref2), V_(ref3) are reference voltages andV_(ee1), V_(ee) are negative power source voltages.

Here, the emitter of emitter follower transistor Q₁₇ forming the emitterfollower output circuit 15 within the D latch circuit is connected withthe pull-down transistor Q₁₉. In addition, the base of pull-downtransistor Q₁₉ is Self-biased, as in the case of the embodiment of FIG.23, with the resistance R₂₂ and receives the charging current of thedifferential capacitance C_(a4) from the side of D latch circuit 21.

Thereby, the D latch circuit 21 of the logic circuit LOG₄ describedabove is capable of executing the high speed data latch operation byimprovement of falling characteristic of the emitter follower outputcircuit 15 provided in the feedback path in such D latch circuit 21.Moreover, since the bipolar transistor Q₂₁ and pull-down transistor Q₁₉are resistive to the α-ray like the bipolar transistor Q₁ and pull-downtransistor Q₃ shown in FIG. 1, the D latch circuit 21 executes highlyreliable holding operation.

FIG. 25 shows an embodiment of the fourth logic circuit to which thepresent invention is applied.

In FIG. 25, the emitter follower output circuits 15, 16, activepull-down circuits 17, 18, bias circuits 19, 20, differentialcapacitances C_(a4), C_(a5) respectively correspond to those ofembodiment shown in FIG. 23 and operate in the same way.

Referring to FIG. 25, the logic circuit LOG₄ is formed by a currentswitching circuit 22 consisting of the bipolar transistors Q₃₀ -Q₃₃ andresistances R₂₉ -R₃₁, emitter follower output circuits 15, 16respectively providing output of the current switching circuit, activepull-down circuits 17, 18, bias circuits 19, 20 and differentialcapacitances C_(a4), C_(a5).

The current switching circuit 22 provides an output respectively to theemitter follower transistors Q₁₇, Q₁₈ in accordance with the digitalinput signals V_(i5), V_(i6) and reference voltages V_(ref4). Thebipolar transistor Q₃₃ receives the predetermined base control voltageV_(b8) (for example, 1.85 V) and forms a constant current source incombination with the resistance R₃₂. As described above, output ofcurrent switching circuit 22 is output by the emitter follower of theemitter follower transistors Q₁₇, Q₁₈ as the digital output signalsV_(o8), V_(o9).

Respective digital output signals V_(o8), V_(o9) are connected byswitching to the power source voltage V_(ee) in the low level side bythe pull-down transistors Q₁₉, Q₂₉.

On the other hand, the differential capacitance C_(a4) differentiatesthe rising edge of digital output signal V_(o9) and supplies thecharging current to the base of pull-down transistor Q₁₉ to temporarilydrive the transistor. Meanwhile, the differential capacitance C_(a5)differentiates the rising edge of digital output signal V_(o10) andsupplies the charging current to the base of pull-down transistor Q₂₀ inorder to temporarily drive the transistor.

Moreover, in the pull-down transistors Q₁₉, Q₂₀, the bias is applieddepending on the load capacitance CL₂ by the feedback of the voltage ofdigital output signals V_(o8) V_(o9) to the base through the resistanceR₂₂, R₂₃ and by the pull-down of the base to the emitter side throughthe resistances R₂₀, R₂₁, and simultaneously the differential outputgenerated when the differential capacitances C_(a4), C_(a5) are chargedrespectively to the digital output signals V_(o8), V_(o9), is enhancedbecause the differential capacitances C_(a4), C_(a5) are charged fromthe side of digital output signals V_(o8), V_(o9) when the these digitaloutput signals are high level.

Thereby, as in the case of the embodiment of FIG. 23 described above,the rising characteristic of the digital output signals V_(o8), V_(o9)output as the complementary signal can be improved.

FIG. 26 indicates an embodiment of the fifth logic circuit to which thepresent invention is applied.

In FIG. 26, the logic circuit LOG₆ is a modification of the embodimentshown FIG. 25 and has the same basic configuration. Explanation will bemade focusing on the difference from FIG. 25. The logic circuit LOG₆includes the complementary signal circuit 23 in which the digital inputsignals V_(i5), V_(i6) are negative input of V_(i5) and thecomplementary signal can be formed by high level or low level of thedigital input signal V_(i5) and the constant current source which isformed by the bipolar transistor Q₃₁ and resistance R₃₁ shown in FIG. 25is designated by IS₂. Moreover, the bias circuits 24, 25 are formed bythe bipolar transistors Q₃₄, Q₃₅ in place of the resistances R₂₂, R₂₃different from the bias circuits 19, 20 of the embodiment shown in FIG.25 and the resistances R₂₀, R₂₁ shown in FIG. 25 are indicated as theresistances R₃₃, R₃₂.

FIG. 27 is a schematic plan view of a large capacity gate arrayintegrated circuit to which the present invention is applied. FIG. 28indicates an embodiment in which the cell of large capacity gate arrayintegrated circuit of FIG. 27 is designated by the logic circuit LOG₁ ofFIG. 14. FIG. 29 indicates a sectional view of the semiconductor chipforming a large capacity gate array integrated circuit shown in FIG. 27sealed with the microchip carrier.

FIG. 30 indicates a sectional view in which a capacitance chip is formedon the carrier shown in FIG. 29. FIG. 31 shows that the microchipcarrier shown in FIG. 29 is mounted on the mounting substrate.

In FIG. 27, a large capacity gate array GA allows, although notparticularly limited, arrangement of many basic cells at the internalside of input/output buffer region I/O formed by the ECL circuit. Eachbasic cell forms the logic circuit, for example, the logic circuit LOG₁shown in FIG. 5, FIG. 8, FIG. 10, FIG. 14, FIG. 16, FIG. 18, FIG. 19,FIG. 22, logic circuits LOG₃ -LOG₆ shown in FIG. 23 - FIG. 26 or logiccircuit LOG₂ shown in FIG. 13. The unused cell cell' of the basic cellsis the capacitive cells in order to prevent malfunction under the steadystate of the pull-down transistor shown in the explanation for FIG. 14.Moreover, in the large capacity gate array integrated circuit GA, thecapacitive element CE is formed between the basic cells Cell in order toprevent malfunction described above. Moreover, the large capacity gatearray GA is provided with a base control voltage generating circuit 24in order to form the base control voltage V_(b1) (V_(b4)) in accordancewith the control voltage V_(c) supplied from the external terminal Incase the logic circuit of each embodiment is formed with the largecapacity gate array integrated circuit GA, many logic circuits can beformed with high density due to the high speed operation characteristicand low power consumption characteristic. In addition, since the wiredlogic can be formed, the application efficiency of the logic circuit canbe remarkably enhanced and a very large scale and high performancesemiconductor integrated circuit device can be obtained efficiently withthe gate array configuration.

In FIG. 28, the left logic circuit is designated as LOG₁, while theright logic circuit as LOG_(1'). Many logic circuits LOG₁ are connectedbetween the power source voltage V_(cc2) and V_(ee3). Moreover, theinductances L₁, L₂ are inductance of CCB bump electrode 29 shown in FIG.29, while the inductances L₃, L₄ are inductance of carrier 25A shown inFIG. 29. Moreover, the capacitance C_(a6) is the capacitance in the chipand is equal to the capacitance of the capacitance element CE and unusedcell cell' shown in FIG. 27 and the capacitance C_(a7) is thecapacitance of carrier 25A shown in FIG. 29, namely the layeredcapacitance 32 or capacitance chip 33 shown in FIG. 29 and FIG. 30.

In FIG. 28, when the digital output signal V_(o1) of logic LOG₁ falls tothe low level from the high level, the pull-down transistor Q₃ istemporarily driven conductively as described previously. In this case,since the emitter follower transistor Q₂ is not yet sufficiently drivennon-conductively, a current I₅ temporarily flows. The power sourcevoltage V_(ee3) changes according to being flow the current I₅ to theinductances L₁, L₂, L₃ and L₄ therefore the pull-down transistor Q₃ isconductively driven due to change of emitter voltage of such pull-downtransistor Q₃ at the steady state of the logic circuit connected to thepower source voltages V_(cc2) and V_(ee3), for example, the logiccircuit LOG_(1'), resulting in possibility of malfunction. However, inthis embodiment, many logic circuits LOG₁ are connected between thepower source voltages V_(cc2) and V_(ee3). The pull-down transistor Q₃under the steady state of the logic circuit LOG₁ is in the very weak ONcondition and the collector current I_(c3) is about 10-100 μA asdescribed previously. Moreover, since the emitters of pull-downtransistors Q₃ of a plurality of logic circuits LOG₁ are connected incommon to the power source voltage V_(ee3), a kind of emitter coupledcircuit is formed. In addition, a small number of pull-down transistorsQ₃ are conductively driven simultaneously in the large capacity gatearray GA. Therefore, since a part of through current I₅ flows onto aloop like the current I₆, the through current flowing into theinductances L₁, L₂, L₃ and L₄ decreases, suppressing variation of powersource voltage V_(ee3). However, since the logic circuit LOG₁ is a superhigh speed circuit, variation of power source voltage V_(ee3) issuppressed and moreover the chip capacitance C_(a6) of the largecapacity gate array integrated circuit GA and the capacitance C_(a7) ofthe carrier 25A shown in FIG. 29 and FIG. 30 are formed in the largecapacity gate array integrated circuit GA of this embodiment. A part ofthe through current I₅ further flows on the loop current I₆, I₇, I₈ andthereby variation of power source voltage V_(ee3) is suppressed.

In FIG. 29, 25 designates microchip carrier; 25A, carrier for mountingchip; 25B, cap; 26, 27, solder; 28, semiconductor chip; 29, CCB(Controlled Collapse Bonding) bump electrode; 30, bump electrode formounting; 31, wiring. The arrow mark at the end of leader line indicatesthe entire part of microchip carrier 25.

The CCB bump electrode 29 is connected to the input/output terminal ofgate array integrated circuit GA and power source terminals of powersource voltages V_(cc) and V_(ee) and is also connected to the bumpelectrode 30 for mounting through the wiring 31.

The microchip carrier 25 of this embodiment forms a layered capacitance32 on the chip mounting carrier 25A in order to prevent malfunctionunder the steady state of the pull-down transistor Q₃.

The CCB bump electrode 29 is described in the Japanese Laid-open PatentNo. 62-249429 and the microchip carrier 25 is described in the JapanesePatent Application No. 62-146397.

In FIG. 30, the basic configuration is the same as that of microchipcarrier 25 shown in FIG. 29. Focusing on difference from the microchipcarrier 25 of FIG. 29, this embodiment prevents malfunction of thepull-down transistor Q₃ under the steady state by not forming thelayered capacitance 32 on the carrier 25A, forming the capacitance chip33 on the carrier 25A and connecting such chip to the wiring 31 throughthe CCB bump electrode 29.

In FIG. 31, 34 designates a mounting substrate; 35 is pin. Manymicrochip carriers 25 are mounted on the substrate 34 as shown in FIG.31. The mounting bump electrode 30 is connected to the pin 35 by aplurality of wired layers (not illustrated) included in the substrate31. The substrate 34 is sealed by the cap, although not illustrated.

As described, in the large capacity gate array integrated circuit GA ofthis embodiment, a plurality of logic circuits LOG₁ are provided betweenthe power source voltages V_(cc2) and V_(ee3) in order to preventmalfunction of the pull-down transistor Q₃ under the steady state due tovariation of the power source voltage V_(ee3). In addition, suchmalfunction can be further prevented by forming capacitance on the chip28 and carrier 25A. Moreover, such malfunction is prevented moresuccessfully by sealing the package of the large capacity gate arrayintegrated circuit GA by the method in which the microchip carrier 25 issealed using the CCB bump electrode 29 having smaller inductance thanthat in the method using the lead wire. Moreover, since the base controlvoltage V_(b1) (V_(b4)) of the bias circuits 4 and 8 and bipolartransistors Q₄ and Q₁₀ can be controlled by changing the control voltageVC shown in FIG. 27. Accordingly, the falling time of the digital outputsignal V_(o1) of the logic circuit LOG₁ can be delayed, for testabilityof the large capacity gate array integrated circuit GA, by controllingthe bias voltage V_(b) (V_(b")) of the pull-down transistor Q₃.

As explained in the embodiments above, in case the present invention isapplied to the large capacity gate array or high speed logic integratedcircuit basically configurated by the NTL circuit or ECL circuit,following effects can be obtained.

(1) The load means can be temporarily short-circuited at the beginningof the low level output and level of output signal can be changedquickly by providing the pull-down transistor which is capacitivelyconnected to the emitter of the transistor receiving the input signal atits base and temporarily shows a large conductance for the low leveloutput.

(2) With the item (1), following effect can be obtained. Namely, goodresponse characteristic of input pulse to the NTL circuit can beobtained and level change at the rising and falling edges is almostequal. The high load driving capability for the NTL circuit can berealized and the output signal in the range from light to heavy load canbe changed quickly by forced removable of capacitance of load means forthe low level output with the pull-down transistor. Since the pull-downtransistor such as NTL circuit shows high impedance under the steadystate, a wired OR circuit which does not easily generateelectromigration can be formed. Employment of the NTL circuit having theconfiguration indicated in the item (1) results in resistivity to theα-ray. With use of the NTL circuit of the configuration described above,the pull-down transistor can be formed without use of a particularelement such as PNP transistor and output signal level can be changedquickly. Since the pull-down transistor can be formed on a singlesemiconductor substrate together with the other circuit elements, highspeed level change of the output signal can be realized by the devicetechnology. Since the NTL circuit is a low amplitude circuit, operationrate can further be improved. Low power consumption of the NTL circuitcan be attained by making large the resistance value of load means andmaking small an operation current of the output emitter followercircuit.

(3) Following effects can be obtained from the items (1) and (2). TheNTL circuit is a low voltage circuit and low power consumption can alsobe realized for the NTL circuit. Since the rising and falling times ofoutput signal are almost equal, simplified delay calculation of the NTLcircuit can be realized easily and dependence on the input waveform ofthe circuit operation rate of NTL circuit can be reduced. The loaddriving capability of the NTL circuit can further be improved byproviding the bias circuit to the base of pull-down transistor of theNTL circuit of the configuration described above. The low powerconsumption can also be realized without interfering the high speedoperation of the NTL circuit.

(4) Following effects can be obtained from the items (1) to (3). Theload driving capability of NTL circuit and falling rate of output signalcan be set adequately by providing adequate capacitance between thepull-down transistor and the transistor receiving the input signal. Theload driving capability of NTL circuit and falling rate of output signalcan be changed by configurating the bias circuit with the transistor tobe controlled with the predetermined base voltage and changing thepredetermined base voltage in accordance with the control voltagesupplied from the external terminal and variation of speed of aplurality of NTL circuits can be suppressed and the testabilityefficiency of the large capacity gate array integrated circuit formed bythe NTL circuit can be improved. The falling rate of output signal canalways be improved effectively irrespective of external environment ofthe load means of the NTL circuit by forming the bias circuit with theresistance which feeds back the output signal. The circuit configurationdescribed above can also be used in the gate circuit or latch circuit byECL circuit in addition to the NTL circuit, resulting in wide rangeoperation principle of the present invention.

(5) From the items (1) to (4), high integration density can be realizedbecause the lower power consumption can be realized without impedinghigh speed operation of the high speed logic integrated circuitincluding the NTL circuit.

While the present invention has been explained based on the embodimentthereof, the present invention is naturally not limited and allowsvarious changes or modifications without departure from the scope of theinvention.

For example, in each embodiment, the high level V_(H) and low levelV_(L) of the digital input signals V_(i1) -V_(i6) and digital outputsignals v_(o1) -V_(o9) can be set to the desired levels. In addition,the power source voltage V_(cc) of circuit may be replaced with apositive source source voltage by setting the power source voltageV_(ee) of circuit to the ground potential and the polarity thereof canalso be inverted by replacing the bipolar transistor with the PNPtransistor. The phase split circuit 1, phase inversion circuit 14, Dlatch circuit 21, current switching circuit 22 and complementary signalcircuit 23 shown in the embodiments of FIG. 5, FIG. 8, FIG. 10-FIG. 14,FIG. 16-FIG. 19 and FIG. 22-FIG. 26 may be formed using the NOS(metal-oxide-semiconductor) transistor, and a speed-up capacitance maybe provided to the resistance R₃ to be provided between the emitter ofbipolar transistor Q₁ forming the phase split circuit 1 and the powersource voltage V_(ee) in the embodiment shown in FIG. 5, FIG. 8, FIG.10-FIG. 14, FIG. 16-FIG. 19 and FIG. 22. Moreover, the embodiments shownin FIG. 10, FIG. 14 and FIG. 22 are capable of providing a desirednumber of fan-in by replacing the bipolar transistor Q₁ forming thephase split circuit 1 with a plurality of transistors connected inparallel. In FIG. 18, the base of pull-down transistor Q₂₃ forming theactive pull-down circuit may be coupled directly to the nodes connectedin common of the resistances R₁₀ and R₁₁ without passing through theresistance R₁₂. In FIG. 19, the diode D₁ provided between the bipolartransistor Q₄ forming the bias circuit 3 and the power source voltageV_(cc) of the circuit may be replaced with a plurality of diodes ortransistors in accordance with the level of constant voltage V_(b3) andthe capacitance C_(a2) may be removed. In addition, combination of apractical circuit configuration and power source voltage in eachembodiment can employ various forms.

In the above explanation, the inventions by the inventors of the presentinvention are applied to the NTL circuit and ECL circuit of the gatearray which is the application field as the background of the presentinvention, but the application field is not limited thereto and thepresent invention can also be applied, for example, to the semiconductorintegrated circuit device such as MPU (microprocessor) of whichapplication field is fixed. The present invention can be applied atleast to the logic circuit including the output emitter follower circuitor various kinds of semiconductor integrated circuit device includingsuch logic circuit.

Another aspect of the present invention will now be discussed As notedpreviously, a conventional logic circuit such as the ECL circuit havingan emitter follower output transistor can have a good risingcharacteristic of the signal by virtue of the low output impedance ofthe emitter follower output transistor. However, since the fallingcharacteristic of the signal is dependent on the time constantdetermined by the capacitance of the load and the resistance of theemitter follower, an LSI for a gate array designed to achieve high-speedoperation and high packaging density inevitably consumes a great amountof power. Therefore, an active pull-down circuit is proposed, forexample, in the gazette of Japanese Laid-Open Patent Publication No.50-134356, which circuit improves the falling characteristic of theoutput signal of an emitter follower output circuit while it keeps lowthe power consumption in the steady state.

There is also an NTL circuit which, different from such as the abovedescribed ECL circuit discriminating between high level and low level byreference to the threshold voltage, has no special logic threshold. Inthe NTL circuit, the signal is amplified by being passed through aplurality of gate circuits. When input/output transfer characteristicsof a plurality of gate circuits cross virtually at the center voltage,an input level higher than the cross point is progressively amplifiedand finally converges to the high level side. In a logic gate circuit, aplurality of logic gates are arranged in multiple stages, and therefore,even if there is fluctuation of the transfer characteristic in the NTLcircuit, level correction is achieved by passing a signal through otherlogic gates having similar fluctuation and, therefore, fluctuation ofthe transfer characteristic practically poses no problem. Such NTLcircuit is described, for example, in Genji Baba, "Modern Dictionary ofElectronic Devices", published by Radio Technology Company on Mar. 20,1976, p. 72. As patents relating to NTL circuits, there are thosedisclosed in Japanese Patent Publication Nos. 42-21132, 42-2294, and 45-32005 published respectively in 1967, 1967 and 1970.

In FIG. 39 is shown a circuit diagram of an example of an ECL circuitprovided with an active pull-down circuit which the present inventorsinvestigated. A transistor Q44 provided for an emitter-follower outputtransistor Q33, in the steady state, is biased to its base-emittervoltage V_(BE) by a bias circuit made up of a resistor R48 and atransistor Q45 so that virtually no current is passed therethrough. Atthe time when the signal is changed and thereby the transistor Q43 isswitched from on state to off state, the transistor Q44 is turned on bya signal with reverse phase transmitted from an emitter-followertransistor Q49 through a capacitor C41, and thereby, the output terminalOUT is quickly discharged.

However, if the signal level should change while the discharging throughthe capacitor C41 is going on, a sufficient falling characteristic ofthe output signal would not be obtained. In order to obtain a sufficientfalling characteristic of the output signal, it is necessary to takesuch measures as to increase the capacitance of the capacitor C41 orincrease the steady-state current flowing through the transistor Q44. Ifsuch measures are taken, however, there arise problems of fluctuation ofthe falling characteristic of the output signal, deterioration in therising characteristic due to undershooting made by the output signal,and increase in the power consumption, which the present inventors foundout through their investigation.

Accordingly, another object of the present invention is to provide alogic circuit which, while keeping power consumption low, achieveshigh-speed signal outputting.

This and other objects and novel features of the present invention willbecome apparent from the description given hereinbelow and theaccompanying drawings.

A representative aspect of this aspect of the present invention is asfollows. An emitter follower output transistor receiving an outputsignal generated by the logic portion is connected in series with anactive pull-down transistor supplied, at the base, with a signal, whichhas reverse phase to that of the input signal supplied to the base ofthe output transistor, through a capacitance element. Between the baseand emitter thereof, there is provided a bias circuit formed of atransistor supplied with a predetermined bias voltage and an emitterresistor, and there is further provided a capacitance element forfeeding back the output signal to the emitter of the transistor formingthe bias circuit.

According to the above-described arrangement, the base potential of theactive pull-down transistor, at the time of the fall of the outputsignal, is lowered by the turning on of the bias transistor caused bythe signal given thereto through the feedback capacitance element, andthereby, excessive action of the active pull-down transistor issuppressed and, hence, occurrence of undershooting of the output can beprevented.

FIG. 34 shows a circuit diagram of an embodiment in which an activepull-down circuit according to the present invention is applied to anNTL circuit. Circuit elements in the diagram are formed on onesemiconductor substrate of single crystalline silicon or the like byknown fabrication technology of bipolar integrated circuit.

The circuit formed of transistors Q41 and Q42 receiving signals frominput terminals IN1 and IN2, a resistor R41 provided for the collectorsconnected with each other of the transistors Q41 and Q42 and a resistorR42 for the emitters connected with each other of the same constitutesan NTL circuit. More specifically, when the signal at the input terminalIN1 or IN2 is high, the transistor Q41 or Q42 is turned on, whereby anoutput signal determined by the ratio of resistance between theresistors R41 and R42 at a level as low as approximately -0.5 V isproduced, and when the signals at the input terminals IN1 and IN2 areboth low, both the transistors Q41 ad Q42 are turned off, whereby asignal b at a level as high as 0 V is generated at the collector. Hencethe NTL circuit constitutes a two-input NOR gate logic circuit.

The collector output signal b of the transistors Q41 and Q42 is suppliedto the base of an emitter-follower output transistor Q43. The transistorQ43 has its emitter connected in series with an active pull-downtransistor Q44. The base of the transistor Q44 is supplied with anoutput signal c with reverse phase obtained from the emitter of thetransistors Q41 and Q42 through a capacitor C41. In order to pass arelatively small idling current through the transistor Q44 in the steadystate, a bias circuit as described below is provided.

Between the base and emitter of the transistor Q44, there are inserted atransistor Q46 receiving a predetermined bias voltage VB2 and itsemitter resistor R44. Further, to the base of the transistor Q44 isconnected the emitter of a transistor Q45 receiving a predetermined biasvoltage VB1. The collector of the transistor Q45 is connected with theground potential point.

Further, in order to prevent occurrence of an undershoot of the outputsignal, there is provided in the present embodiment a capacitor C42allowing the output signal to be fed back to the emitter of the biastransistor Q42.

For example, when the signals at the input terminals IN1 and IN2 areboth at low level and hence both the transistors Q41 and Q42 are in theoff state, their collector output becomes as high as 0 V, the groundpotential. This high level is subjected to a level shift through thebase and emitter of the emitter-follower output transistor Q43, and as aresult, the potential at the output terminal OUT is brought to a levelas high as approximately -0.8 V.

From this state, if the signal a at the input terminal IN1 changes fromlow level to high level, the output signal b changes from high level tolow level. As a result, the output transistor Q43 is virtually put intoan off state and the output signal c with reverse phase obtained fromthe emitter of the transistor Q41 is changed from low level to highlevel, and this change to high level causes, through the capacitor C41,the base potential of the transistor Q44 to he temporarily held high.Thereby, the transistor Q44 is turned on and the output signal d of theoutput terminal OUT is quickly changed from high level to low level.Then, this change causes, through the capacitor C42, the emitterpotential of the transistor Q46 to go to low level in a manner as if itwere an A.C. wave. Thereby, a collector current flows through thetransistor Q46, and this causes the base potential of the transistor Q44to lower. In other words, the high potential transmitted from thecapacitor C41 to the base of the transistor Q44 in accordance with thechange of the signal c to high level is repressed by the low levelsignal fed back through the capacitor C42. Thus, an undershoot of theoutput signal d is prevented from occurring.

Since the occurrence of the undershoot of the output signal d can beprevented as described above, it becomes possible to set the capacitanceof the capacitor C41 to a considerably high value for improvement of thefalling characteristic to the low level of the signal d and yet toprevent occurrence of the undershoot of the signal d by the feedbackcontrol with the capacitor C42. This means in substance that the risingcharacteristic of the base potential of the transistor Q44 can beprevented from deteriorating.

For the capacitor C41, a relatively large capacitance value can be setup without substantially increasing the layout area by utilizing thebase take-out electrode of the transistor Q42. Also for the feedbackcapacitor C42, the take-out electrode of the emitter of the transistorQ46 and others can be utilized so as to achieve reduction in the layoutarea.

The operating voltage supply for the output circuit through which arelatively large current flows can be arranged separate from thai forthe input logic portion through which a relatively small current flows.That is, the separation is made such that the power source lines for theinput logic portion are comprised of Vcc and -VEE, while those for theoutput portion are comprised of Vcc' and -VEE'. By virtue of sucharrangement, the level margin of the signal can be prevented from beingdegraded by noises which are produced in the power line Vcc' or -VEE'during the operation of the output circuit and transmitted to the inputlogic portion.

FIG. 35 shows a circuit diagram of another embodiment in which an activepull-down circuit according to the present invention is applied to anNTL circuit.

In this embodiment, the output signal d is fed back to the base of thetransistor Q45 through a capacitor C43. Though not limitative, the biasvoltages supplied to the bases of the transistors Q46 and Q45 areprovided by a series circuit of diodes D41 and D42 and a resistor R45.More specifically, the base of the transistor Q46 is supplied with aforward voltage V_(F) across the diode D41 The base of the transistorQ45 is supplied with a forward voltage 2 V_(F) across the diodes D41 andD42. The resistor R45 forms a constant bias current to be passed throughthe diodes D41 and D42. That is, a constant current -(VEE-2 V_(F))/R45is passed through the resistor R45 and this constant current is passedthrough the transistors Q45 and Q46.

In the described arrangement, the circuit producing the bias voltagesVB1 and VB2 can be formed of such simple circuit elements as theresistor R45 and the diodes D41 and D42.

The feedback capacitor C43 improves the rising characteristic of theoutput signal d. More specifically, when the output signal d goes fromlow level to high level, this change in level acts through the capacitorC43 on the transistor Q45 to raise its base potential. Thereby, theaction of the capacitor C41 when the level of the signal c with reversephase changes from high level to low level exerted on the transistor Q4to lower its base potential is repressed. Generally, when the basepotential of the transistor Q44 goes lower than that in the steadystate, the transistor Q44 is turned on after the steady state has beenrestored. Hence, the response characteristic when the signal c isquickly changed from low level to high level is deteriorated. In thepresent embodiment, by virtue of the feedback signal transmitted throughthe capacitor C43, the base potential of the transistor Q44, when theoutput signal d goes high, is prevented from going lower than the biaspoint by the action exerted thereon through the capacitor C41.

Thus, in the present embodiment, the response characteristics to theswitched input are greatly improved by means of the feedback capacitorsC42 and C43.

FIG. 36 shows a circuit diagram of a further embodiment in which anactive pull-down circuit according to the present invention is appliedto an NTL circuit.

When an active pull-down circuit is used, it is required to provide acircuit to generate a bias voltage VB1 allowing an idling current toflow through the pull-down transistor Q44 as described above. The biascircuit in the present embodiment is structured of diodes D41 and D42and a resistor R45 in series connection, the same as that used in theabove described embodiment.

More specifically, a forward voltage 2 V_(F) produced by the diodes D41and D42 is supplied to the base of the transistor Q45. Thus, thetransistor Q45 and the diodes D41 and D42 are arranged in a currentmirror configuration and it is made possible to allow a constant currentprovided by the resistor R45 to flow through the transistors Q45 andQ44. With such arrangement, the bias circuit when the active pull-downcircuit is used can be simply structured.

FIG. 37 shows an embodiment in which an active pull-down circuitaccording to the present invention is applied to an ECL circuit.

In the ECL circuit, an output signal and a signal with reverse phase areobtained from a differential transistor circuit. Accordingly, the signalwith reverse phase to be transmitted to the base of the active pull-downtransistor Q44 is obtained from the emitter of an emitter followeroutput transistor Q49 receiving the collector output of an invertingoutput transistor Q47 supplied with a reference voltage VBB.

In the present case, the load resistor in the differential transistorcircuit is divided into two, RC2 and RC3, whereby the level of thesignal transmitted to the base of the active pull-down transistor Q44through the capacitor C41 is set up, and the response characteristic ofthe base of the active pull-down Q44 can be optimally set up. Thus, theundershoot produced by an excessively high level of the signaltransmitted to the base of the transistor Q44 through the capacitor C41and excessive reverse biasing of the transistor Q44 at the timing of therise of the output signal can be prevented from occurring.

The bias circuit of the present invention is structured of the diodesD41 and D42 and the resistor R45 connected in series the same as thebias circuit used in the above described embodiment. More specifically,a forward voltage 2 V_(F) produced by the diodes D41 and D42 is suppliedto the base of the transistor Q45. Thus, the transistor Q45 and thediodes D41 and D42 are arranged in a current mirror configuration and itis made possible to allow a constant current provided by the resistorR45 to flow through the transistors Q41 and Q42. With such arrangement,the bias circuit when the active pull-down circuit is used can be simplyprovided. Instead of the above described arrangement, the bias circuitmay be such that supplies a predetermined bias voltage VB1 to the baseof the transistor Q45.

FIG. 38 shows a circuit diagram of another embodiment of an activepull-down circuit according to the resent invention applied to an NTLcircuit.

In this embodiment, a diode D43 is used as a variable capacitanceelement for having the output signal fed back to the base of the activepull-down transistor Q44. In this arrangement, when the input signal IN1goes high, the diode D43 is reversely biased and its capacitance valueis decreased. Thereby, the portion escaping through the diode D43 out ofthe pulse current to be supplied to the base of the transistor Q44through the capacitor C41 can be reduced. In contrast with this, whenthe input signal IN1 goes low, the lowering of the potential at the baseof the transistor Q44 by the action exerted thereon through thecapacitor C41 can be compensated for by the rise of the output signal atthe output terminal OUT fed back thereto through the diode D43. At thistime, the reverse bias voltage is lowered and, proportionally to it, thecapacitance is increased, and hence a great fed back quantity can beobtained. In short, the diode D43 has the same function as that of thecapacitor C43 in FIG. 2. In the case of the circuit of FIG. 35, thecapacitor C43 and the capacitor C41 are separated by the transistor Q45and no problem is posed. In the case of the present embodiment, sincethe capacitor C41 and the feedback capacitance element are in seriesconnection, the signal to be transmitted to the base of the transistorQ44 through the capacitor C41 may escape to the output terminal sidethrough the feedback capacitance element, but this escaping can beprevented by the use of the diode D43 as a variable capacitance element.

The transistor Q45 receiving the bias voltage VB1 is for allowing a biascurrent to flow through the transistor Q44 and the transistor Q45', andthe resistor R44', provided for its emitter are for producing a biasvoltage provided to the diode D43 as a reverse bias voltage so that itmay act as the capacitance element.

Functional effects obtained from the above described embodiments are asfollows:

(1) An emitter follower output transistor receiving an output signalgenerated by the logic portion is connected in series with an activepull-down transistor supplied, at the base, with a signal, which hasreverse phase to that of the input signal supplied to the base of theoutput transistor, through a capacitance element. Between the base andemitter thereof, there is provided a bias circuit formed of a transistorsupplied with a predetermined bias voltage and an emitter resistor, andthere is further provided a capacitance element for feeding back theoutput signal to the emitter of the transistor forming the bias circuit.According to the above described arrangement, the base potential of theactive pull-down transistor, at the time of the fall of the outputsignal, is lowered by the turning on of the bias transistor caused bythe signal given thereto through the feedback capacitance element, andthereby, excessive action of the active pull-down transistor issuppressed and, hence, occurrence of undershooting of the output can beprevented while the falling characteristic of the output signal isimproved.

(2) An emitter follower output transistor receiving an output signalgenerated by the logic portion is connected in series with an activepull-down transistor supplied, at the base, with a signal, which hasreverse phase to that of the input signal supplied to the base of theoutput transistor, through a capacitance element. As a bias circuitprovided for the base of the active pull-down transistor, a bias currentsupplying transistor whose emitter is connected with the base of theactive pull-down transistor, two forward diodes disposed between thebase of the bias current supplying transistor and the emitter of theactive pull-down transistor, and a resistance element allowing a biascurrent to flow through the forward diodes. Thus, an effect is obtainedthat configuration of the circuit can be simplified.

(3) When the logic circuit is constituted of an ECL circuit, the loadresistor for forming the reverse output signal in the differentialtransistor circuit is divided in two, whereby the signal level suppliedto the base of the active pull-down transistor is made adjustable. Thus,an effect is obtained that setting for optimal operation of the activepull-down transistor can be achieved with a simple circuitconfiguration.

While the invention made by the present inventor has been describedconcretely as related to some embodiments, it is apparent that thepresent invention is not limited to the above described embodiments butvarious modifications may be made without departing from the spirit ofthe present invention. For example, in the circuits of embodiments inFIG. 34 and FIG. 35, the input logic portion may be replaced by an ECLcircuit. The load resistor RL provided for the output terminal OUT inthe circuit for each embodiment is just for generally representing aload and it does not mean that such a resistance element is connected.The power supply lines and operating voltages of the logic portion andthe output portion may be separated according to the need. The number ofthe input terminals may he increased or decreased according to the need.

The present invention can he widely applied to logic circuits comprisedof a high-speed logic portion such as NTL and ECL and an activepull-down circuit.

We claim:
 1. A logic circuit comprising:an input terminal; an outputterminal; a logic portion which receives an input signal supplied tosaid input terminal, and which generates first and second output signalsin accordance with said input signal; an output portion including: afirst bipolar transistor having its collector-emitter path coupled tosaid output terminal and its base coupled to receive said first outputsignal, and a second bipolar transistor having its collector-emitterpath coupled to said output terminal, and having a base; a firstcapacitance element having a first electrode coupled to receive saidsecond output signal and a second electrode coupled to said base of saidsecond bipolar transistor, wherein a signal supplied to said base ofsaid second bipolar transistor through said first capacitance elementhas a phase reverse to that of said first output signal supplied to thebase of said first bipolar transistor; a bias circuit having a portionthereof coupled between the base and emitter of said second bipolartransistor and including a third bipolar transistor; and a secondcapacitance element, coupled between the output terminal and an emitterof said third bipolar transistor, for feeding back a voltage change ofan output signal on said output terminal to said emitter of said thirdbipolar transistor.
 2. A logic circuit according to claim 1, whereinsaid logic portion is comprised of an NTL circuit which includes aninput bipolar transistor having it collector coupled to a collector loadelement, its emitter coupled to an emitter lead element and to the firstelectrode of said first capacitance element, and its base coupled tosaid input terminal, wherein the first output signal is generated formthe collector of said input bipolar transistor, and wherein the secondoutput signal is generated from the emitter of said input bipolartransistor.
 3. A logic circuit according to claim 1, wherein said logicportion is comprised of an ECL circuit which includes:an input bipolartransistor having a collector coupled to a first collector load element,a base coupled to said input terminal, and having an emitter, areference bipolar transistor having a collector coupled to second andthird collector load elements, its base coupled to receive a referencevoltage, and its emitter coupled to the emitter of said input bipolartransistor to form a common emitter, and a current source coupled tosaid common emitter, wherein the first output signal is generated fromthe collector of said input bipolar transistor, and wherein the secondoutput signal is generated from a connection point between said secondand third collector load elements.
 4. A logic circuit according to claim1, wherein said third bipolar transistor has its collector coupled withthe base of said second bipolar transistor, its emitter coupled with oneend of said second capacitance element, and its base applied with apredetermined bias voltage.
 5. A logic circuit according to claim 4,wherein said bias circuit further includes an emitter resistor coupledwith the emitter of said third bipolar transistor.
 6. A logic circuitcomprising:an input terminal; an output terminal; a logic portion whichreceives an input signal supplied to said input terminal, and whichgenerates first and second output signals in accordance with said inputsignal; an output portion including: a first bipolar transistor havingits collector-emitter path coupled to said output terminal and its basecoupled to receive said first output signal, and a second bipolartransistor having its collector-emitter path coupled to said outputterminal, and having a base; a first capacitance element having a firstelectrode coupled to receive said second output signal and a secondelectrode coupled to said base of said second bipolar transistor,wherein a signal supplied to said base of said second bipolar transistorthrough said first capacitance element, has a phase reverse to that ofsaid first output signal supplied to the base of said first bipolartransistor; a resistance element coupled between the base and emitter ofsaid second bipolar transistor; a third bipolar transistor having itsemitter coupled with the base of said second bipolar transistor forsupplying a bias current to the base of said second bipolar transistor;and forward diodes coupled between the base of said third bipolartransistor and the emitter of said second bipolar transistor; aresistance element coupled to said forward diodes for forming areference current through said forward diodes, wherein said bias currentis formed in accordance with the reference current.
 7. A logic circuitaccording to claim 6, wherein said logic portion is comprised of an NTLcircuit which includes an input bipolar transistor having its collectorcoupled to a collector load element, its emitter coupled to an emitterload element and to the first electrode of said first capacitanceelement and its base coupled to said input terminal, wherein the firstoutput signal is generated from the collector of said input bipolartransistor, and wherein the second output signal is generated from theemitter of said input bipolar transistor.
 8. A logic circuit accordingto claim 6, wherein said logic portion is comprised of an ECL circuitwhich includes:an input bipolar transistor having a collector coupled tosecond and third collector load elements, its base coupled to receive areference voltage, and its emitter coupled to the emitter of said inputbipolar transistor to form a common emitter, and a current sourcecoupled to said common emitter, wherein the first output signal isgenerated from the collector of said input bipolar transistor, andwherein the second output signal is generated from a connection pointbetween said second and third collector load elements.
 9. Asemiconductor integrated circuit device comprising:a first inputterminal; first and second voltage supply terminals; an output terminal;a first input NPN bipolar transistor having a base coupled to the firstinput terminal, a collector and an emitter; a first load element coupledbetween the first voltage supply terminal and the collector of the firstinput NPN bipolar transistor; a second load element coupled between theemitter of the first input NPN bipolar transistor and the second voltagesupply terminal; a first output NPN bipolar transistor having a basecoupled to the collector of the first input NPN bipolar transistor, anda collector-emitter path coupled between the first voltage supplyterminal and the output terminal; a second output NPN bipolar transistorhaving a base and collector-emitter path coupled between the outputterminal and the first voltage supply terminal; a capacitor elementhaving a first electrode coupled to the emitter of the first input NPNbipolar transistor and a second electrode coupled to the base of thesecond output NPN bipolar transistor; and a bias circuit for biasing thebase of the second output NPN bipolar transistor, the bias circuitincluding:a first bipolar transistor having a collector-emitter pathcoupled between the first voltage supply terminal, the base of thesecond NPN bipolar transistor and a base, a resistor element coupledbetween the first voltage supply terminal and the base of the firstbipolar transistor, and a pair of diodes coupled in series between thebase of the first bipolar transistor and the second voltage supplyterminal so that the first bipolar transistor and the pair of diodes arearranged in a current mirror configuration.
 10. A semiconductorintegrated circuit device according to claim 9, wherein the bias circuitfurther includes:a resistor element coupled between the base of thesecond output NPN bipolar transistor and the second voltage supplyterminal.
 11. A semiconductor integrated circuit device according toclaim 10, further comprising:a capacitor coupled between the outputterminal and the base of the first bipolar transistor.
 12. Asemiconductor integrated circuit device according to claim 9, whereinthe bias circuit includes:a second bipolar transistor having acollector-emitter path coupled between the base of the second output NPNbipolar transistor and the second voltage supply terminal, and having abase coupled to a connection point of the pair of diodes.
 13. Asemiconductor integrated circuit device according to claim 12, furthercomprising:a capacitor coupled between the output terminal and the baseof the first bipolar transistor bipolar transistor in the bias circuit.14. A semiconductor integrated circuit device according to claim 13,wherein the bias circuit further includes:an emitter resistor coupledbetween an emitter of the second bipolar transistor in the bias circuitand the second voltage supply terminal.
 15. A semiconductor integratedcircuit device according to claim 14, further comprising:a capacitorcoupled between the output terminal and the emitter of the secondbipolar transistor in the bias circuit.
 16. A semiconductor integratedcircuit device according to claim 9, wherein the first and second loadelements are resistor elements.
 17. A semiconductor integrated circuitdevice according to claim 9, further comprising:a second input terminal;and a second input NPN bipolar transistor having a base coupled to thesecond input terminal, a collector coupled to the collector of the firstinput NPN bipolar transistor and an emitter coupled to the emitter ofthe first input NPN bipolar transistor.
 18. A semiconductor integratedcircuit device according to claim 9, wherein the capacitor elementincludes a dielectric film between the first and second electrodesthereof.
 19. A logic circuit according to claim 1, wherein the thirdbipolar transistor is coupled between the base and the emitter of thesecond bipolar transistor.